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QT118H-IS

型号：

QT118H-IS

描述：

电荷转移触摸传感器[ CHARGE-TRANSFER TOUCH SENSOR ]

品牌：

ETC[ ETC ]

页数：

13 页

PDF大小：

431 K

下载PDF手册
lQ  
QProx™ QT118H  
OUCH  
C
HARGE-TRANSFER  
T
S
ENSOR  
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Less expensive than many mechanical switches  
Projects a ‘touch button’ through any dielectric  
Turns small objects into intrinsic touch sensors  
100% autocal for life - no adjustments required  
Only one external part required - a 1¢ capacitor  
Vdd  
Out  
1
2
3
4
8
7
6
5
Vss  
Piezo sounder direct drive for ‘tactile’ click feedback  
LED drive for visual feedback  
Sns2  
Sns1  
Gain  
3V 20µA single supply operation  
Opt1  
Opt2  
Toggle mode for on/off control (strap option)  
10s or 60s auto-recalibration timeout (strap option)  
Pulse output mode (strap option)  
Gain settings in 3 discrete levels  
Simple 2-wire operation possible  
HeartBeat™ health indicator on output  
APPLICATIONS -  
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Light switches  
Industrial panels  
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Appliance control  
Security systems  
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Access systems  
Pointing devices  
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Elevator buttons  
Toys & games  
The QT118H charge-transfer (“QT’”) touch sensor is a self-contained digital IC capable of detecting near-proximity or touch. It will  
project a sense field through almost any dielectric, like glass, plastic, stone, ceramic, and most kinds of wood. It can also turn  
small metal-bearing objects into intrinsic sensors, making them respond to proximity or touch. This capability coupled with its  
ability to self calibrate continuously can lead to entirely new product concepts.  
It is designed specifically for human interfaces, like control panels, appliances, toys, lighting controls, or anywhere a mechanical  
switch or button may be found; it may also be used for some material sensing and control applications provided that the presence  
duration of objects does not exceed the recalibration timeout interval.  
The IC requires only a common inexpensive capacitor in order to function. A bare piezo beeper can be connected to create a  
‘tactile’ feedback clicking sound; the beeper itself then doubles as the required external capacitor, and it can also become the  
sensing electrode. An LED can also be added to provide visual sensing indication. With a second inexpensive capacitor the device  
can operated in 2-wire mode, where both power and signal traverse the same wire pair to a host. This mode allows the sensor to  
be wired to a controller with only a twisted pair over a long distances.  
Power consumption is under 20µA in most applications, allowing operation from Lithium cells for many years. In most cases the  
power supply need only be minimally regulated.  
The IC’s RISC core employs signal processing techniques pioneered by Quantum; these are specifically designed to make the  
device survive real-world challenges, such as ‘stuck sensor’ conditions and signal drift. Even sensitivity is digitally determined and  
remains constant in the face of large variations in sample capacitor Cs and electrode Cx. No external switches, opamps, or other  
analog components aside from Cs are usually required.  
The device includes several user-selectable built in features. One, toggle mode, permits on/off touch control, for example for light  
switch replacement. Another makes the sensor output a pulse instead of a DC level, which allows the device to 'talk' over the  
power rail, permitting a simple 2-wire interface. The Quantum-pioneered HeartBeat™ signal is also included, allowing a host  
controller to monitor the health of the QT118H continuously if desired. By using the charge transfer principle, the IC delivers a  
level of performance clearly superior to older technologies in a highly cost-effective package.  
AVAILABLE OPTIONS  
TA  
00C to +700C  
-400C to +850C  
SOIC  
8-PIN DIP  
QT118H-D  
-
QT118H-S  
QT118H-IS  
lq  
©1999-2000 Quantum Research Group  
R1.03 / 0302  
Figure 1-1 Standard mode options  
1 - OVERVIEW  
The QT118H is a digital burst mode charge-transfer (QT)  
sensor designed specifically for touch controls; it includes all  
hardware and signal processing functions necessary to  
provide stable sensing under a wide variety of changing  
conditions. Only a single low cost, non-critical capacitor is  
required for operation.  
+2.5 to 5  
SENSING  
ELECTRODE  
1
Vdd  
2
3
4
7
5
6
OUT  
SNS2  
GAIN  
SNS1  
Figure 1-1 shows the basic QT118H circuit using the device,  
a conventional output drive and power supply  
with  
Cs  
connections. Figure 1-2 shows a second configuration using  
a common power/signal rail which can be a long twisted pair  
from a controller; this configuration uses the built-in pulse  
mode to transmit the output state to the host controller.  
OPT1  
OPT2  
2nF - 500nF  
Cx  
Vss  
OUTPUT = DC  
TIMEOUT = 10 Secs  
TOGGLE = OFF  
GAIN = HIGH  
1.1 BASIC OPERATION  
8
The QT118H employs short, ultra-low duty cycle bursts of  
charge-transfer cycles to acquire its signal. Burst mode  
permits power consumption in the low microamp range,  
dramatically reduces RF emissions, lowers susceptibility to  
EMI, and yet permits excellent response time. Internally the  
signals are digitally processed to reject impulse noise, using  
Cs is thus non-critical; as it drifts with temperature, the  
threshold algorithm compensates for the drift automatically.  
A simple circuit variation is to replace Cs with a bare piezo  
sounder (Section 2), which is merely another type of  
capacitor, albeit with a large thermal drift coefficient. If Cpiezo  
is in the proper range, no other external component is  
required. If Cpiezo is too small, it can simply be ‘topped up’  
with an inexpensive ceramic capacitor connected in parallel  
with it. The QT118H drives a 4kHz signal across SNS1 and  
SNS2 to make the piezo (if installed) sound a short tone for  
75ms immediately after detection, to act as an audible  
confirmation.  
a 'consensus' filter which requires four consecutive  
confirmations of a detection before the output is activated.  
The QT switches and charge measurement hardware  
functions are all internal to the QT118H (Figure 1-3). A 14-bit  
single-slope switched capacitor ADC includes both the  
required QT charge and transfer switches in a configuration  
that provides direct ADC conversion. The ADC is designed to  
dynamically optimize the QT burst length according to the  
rate of charge buildup on Cs, which in turn depends on the  
values of Cs, Cx, and Vdd. Vdd is used as the charge  
reference voltage. Larger values of Cx cause the charge  
transferred into Cs to rise more rapidly, reducing available  
resolution; as a minimum resolution is required for proper  
operation, this can result in dramatically reduced apparent  
gain. Conversely, larger values of Cs reduce the rise of  
differential voltage across it, increasing available resolution  
by permitting longer QT bursts. The value of Cs can thus be  
increased to allow larger values of Cx to be tolerated  
(Figures 4-1, 4-2, 4-3 in Specifications, rear).  
Option pins allow the selection or alteration of several  
special features and sensitivity.  
1.2 ELECTRODE DRIVE  
The internal ADC treats Cs as a floating transfer capacitor;  
as a direct result, the sense electrode can be connected to  
either SNS1 or SNS2 with no performance difference. In both  
cases the rule Cs >> Cx must be observed for proper  
operation. The polarity of the charge buildup across Cs  
during a burst is the same in either case.  
It is possible to connect separate Cx and Cx’ loads to SNS1  
and SNS2 simultaneously, although the result is no different  
than if the loads were connected together at SNS1 (or  
SNS2). It is important to limit the  
The IC is highly tolerant of changes in Cs since it computes  
the threshold level ratiometrically with respect to absolute  
load, and does so dynamically at all times.  
amount of stray capacitance on  
Figure 1-2 2-wire operation, self-powered  
both terminals, especially if the load  
Cx is already large, for example by  
minimizing trace lengths and widths  
so as not to exceed the Cx load  
specification and to allow for a  
larger sensing electrode size if so  
desired.  
The PCB traces, wiring, and any  
components associated with or in  
contact with SNS1 and SNS2 will  
become touch sensitive and should  
be treated with caution to limit the  
touch area to the desired location.  
Multiple touch electrodes can be  
used, for example to create a  
control button on both sides of an  
lq  
1
object, however it is impossible for the  
sensor to distinguish between the two  
touch areas.  
Figure 1-3 Internal Switching & Timing  
ELECTRODE  
Result  
SNS2  
1.3 ELECTRODE DESIGN  
1.3.1 ELECTRODE  
GEOMETRY AND  
S
IZE  
There is no restriction on the shape of  
the electrode; in most cases common  
sense and a little experimentation can  
result in a good electrode design. The  
QT118H will operate equally well with  
long, thin electrodes as with round or  
square ones; even random shapes are  
acceptable. The electrode can also be  
Cs  
Start  
Cx  
Done  
SNS1  
a
3-dimensional surface or object.  
Sensitivity is related to electrode  
surface area, orientation with respect  
to the object being sensed, object  
composition, and the ground coupling  
quality of both the sensor circuit and  
the sensed object.  
C harge  
Amp  
Even when battery powered, just the physical size of the  
PCB and the object into which the electronics is embedded  
will generally be enough to couple a few picofarads back to  
local earth.  
If a relatively large electrode surface is desired, and if tests  
show that the electrode has more capacitance than the  
QT118H can tolerate, the electrode can be made into a  
1.3.3 VIRTUAL  
C
APACITIVE  
GROUNDS  
When detecting human contact (e.g. a fingertip), grounding  
of the person is never required. The human body naturally  
has several hundred picofarads of ‘free space’ capacitance  
to the local environment (Cx3 in Figure 1-5), which is more  
than two orders of magnitude greater than that required to  
create a return path to the QT118H via earth. The QT118H's  
PCB however can be physically quite small, so there may be  
little ‘free space’ coupling (Cx1 in Figure 1-5) between it and  
the environment to complete the return path. If the QT118H  
circuit ground cannot be earth grounded by wire, for example  
via the supply connections, then a ‘virtual capacitive ground’  
may be required to increase return coupling.  
A ‘virtual capacitive ground’ can be created by connecting  
the QT118H’s own circuit ground to:  
sparse mesh (Figure 1-4) having lower Cx than a solid plane.  
Sensitivity may even remain the same, as the sensor will be  
operating in a lower region of the gain curves.  
(1) A nearby piece of metal or metallized housing;  
Figure 1-5 Kirchoff's Current Law  
1.3.2 KIRCHOFF  
’S  
CURRENT  
LAW  
Like all capacitance sensors, the QT118H relies on Kirchoff’s  
Current Law (Figure 1-5) to detect the change in capacitance  
of the electrode. This law as applied to capacitive sensing  
requires that the sensor’s field current must complete a loop,  
returning back to its source in order for capacitance to be  
sensed. Although most designers relate to Kirchoff’s law with  
regard to hardwired circuits, it applies equally to capacitive  
field flows. By implication it requires that the signal ground  
and the target object must both be coupled together in some  
manner for a capacitive sensor to operate properly. Note that  
there is no need to provide actual hardwired ground  
connections; capacitive coupling to ground (Cx1) is always  
sufficient, even if the coupling might seem very tenuous. For  
example, powering the sensor via an isolated transformer  
will provide ample ground coupling, since there is  
capacitance between the windings and/or the transformer  
core, and from the power wiring itself directly to 'local earth'.  
C
X2  
Sense E lectrode  
SENSOR  
C
X1  
C
X3  
Surro und ing e nviro nm ent  
lq  
2
1.3.5 SENSITIVITY  
Figure 1-6 Shielding Against Fringe Fields  
The QT118H can be set for one of 3 gain levels using option  
pin 5 (Table 1-1). This sensitivity change is made by altering  
the internal numerical threshold level required for  
a
detection. Note that sensitivity is also a function of other  
things: like the value of Cs, electrode size, shape, and  
orientation, the composition and aspect of the object to be  
sensed, the thickness and composition of any overlaying  
panel material, and the degree of ground coupling of both  
sensor and object.  
Sense  
wire  
Sense  
wire  
1.3.5.1 Increasing Sensitivity  
In some cases it may be desirable to increase sensitivity  
further, for example when using the sensor with very thick  
panels having a low dielectric constant.  
Sensitivity can often be increased by using a bigger  
electrode, reducing panel thickness, or altering panel  
composition. Increasing electrode size can have diminishing  
returns, as high values of Cx will reduce sensor gain  
(Figures 4-1, 4-2). The value of Cs also has a dramatic  
effect on sensitivity, and this can be increased in value (up to  
a limit). Also, increasing the electrode's surface area will not  
substantially increase touch sensitivity if its diameter is  
already much larger in surface area than the object being  
detected. The panel or other intervening material can be  
made thinner, but again there are diminishing rewards for  
doing so. Panel material can also be changed to one having  
a higher dielectric constant, which will help propagate the  
field through to the front. Locally adding some conductive  
material to the panel (conductive materials essentially have  
an infinite dielectric constant) will also help; for example,  
adding carbon or metal fibers to a plastic panel will greatly  
increase frontal field strength, even if the fiber density is too  
low to make the plastic bulk-conductive.  
Unshielded  
Electrode  
Shielded  
Electrode  
(2) A floating conductive ground plane;  
(3) A nail driven into a wall when used with small  
electrodes;  
(4) A larger electronic device (to which its output might be  
connected anyway).  
Free-floating ground planes such as metal foils should  
maximize exposed surface area in a flat plane if possible. A  
square of metal foil will have little effect if it is rolled up or  
crumpled into a ball. Virtual ground planes are more  
effective and can be made smaller if they are physically  
bonded to other surfaces, for example a wall or floor.  
Table 1-1 Gain Setting Strap Options  
1.3.4 FIELD  
S
HAPING  
Gain  
High  
Tie Pin 5 to:  
Floating  
Pin 6  
The electrode can be prevented from sensing in undesired  
directions with the assistance of metal shielding connected  
to circuit ground (Figure 1-6). For example, on flat surfaces,  
the field can spread laterally and create a larger touch area  
than desired. To stop field spreading, it is only necessary to  
surround the touch electrode on all sides with a ring of metal  
connected to circuit ground; the ring can be on the same or  
opposite side from the electrode. The ring will kill field  
spreading from that point outwards.  
Medium  
Low  
Pin 7  
1.3.5.2 Decreasing Sensitivity  
In some cases the QT118H may be too sensitive, even on  
low gain. In this case gain can be lowered further by a  
number of strategies: making the electrode smaller, making  
the electrode into  
a
sparse mesh using  
a
space-to-conductor ratio (Figure 1-4), or by decreasing Cs.  
high  
If one side of the panel to which the electrode is fixed has  
moving traffic near it, these objects can cause  
inadvertent detections. This is called ‘walk-by’  
and is caused by the fact that the fields radiate  
from either surface of the electrode equally well.  
Again, shielding in the form of a metal sheet or  
foil connected to circuit ground will prevent  
walk-by; putting a small air gap between the  
grounded shield and the electrode will keep the  
Figure 2-1 Drift Compensation  
Signal  
Hysteresis  
Threshold  
value of Cx lower and is encouraged. In the case  
of the QT118H, the sensitivity is low enough that  
'walk-by' should not be a concern if the product  
has more than a few millimeters of internal air  
gap; if the product is very thin and contact with  
the product's back is a concern, then some form  
of rear shielding may be required.  
Reference  
Output  
lq  
3
The QT118H employs a hysteresis dropout below the  
threshold level of 17% of the delta between the reference  
and threshold levels.  
2 - QT118H SPECIFICS  
2.1 SIGNAL PROCESSING  
The QT118H processes all signals using 16 bit precision,  
using a number of algorithms pioneered by Quantum. The  
algorithms are specifically designed to provide for high  
survivability in the face of all kinds of adverse environmental  
changes.  
2.1.3 MAX  
ON-DURATION  
If an object or material obstructs the sense pad the signal  
may rise enough to create a detection, preventing further  
operation. To prevent this, the sensor includes a timer which  
monitors detections. If a detection exceeds the timer setting,  
the timer causes the sensor to perform a full recalibration.  
This is known as the Max On-Duration feature.  
2.1.1 DRIFT  
COMPENSATION  
ALGORITHM  
Signal drift can occur because of changes in Cx and Cs over  
time. It is crucial that drift be compensated for, otherwise  
false detections, non-detections, and sensitivity shifts will  
follow.  
After the Max On-Duration interval, the sensor will once  
again function normally, even if partially or fully obstructed,  
to the best of its ability given electrode conditions. There are  
two timeout durations available via strap option: 10 and 60  
seconds.  
Drift compensation (Figure 2-1) is performed by making the  
reference level track the raw signal at a slow rate, but only  
while there is no detection in effect. The rate of adjustment  
must be performed slowly, otherwise legitimate detections  
could be ignored. The QT118H drift compensates using a  
slew-rate limited change to the reference level; the threshold  
and hysteresis values are slaved to this reference.  
Table 2-1 Output Mode Strap Options  
Tie  
Pin 3 to:  
Tie  
Pin 4 to:  
Max On-  
Duration  
DC Out  
DC Out  
Toggle  
Pulse  
Vdd  
Vdd  
Gnd  
Gnd  
Vdd  
Gnd  
Gnd  
Vdd  
10s  
60s  
10s  
10s  
Once an object is sensed, the drift compensation  
mechanism ceases since the signal is legitimately high, and  
therefore should not cause the reference level to change.  
The QT118H's drift compensation is 'asymmetric': the  
reference level drift-compensates in one direction faster than  
it does in the other. Specifically, it compensates faster for  
decreasing signals than for increasing signals. Increasing  
signals should not be compensated for quickly, since an  
approaching finger could be compensated for partially or  
entirely before even touching the sense pad. However, an  
obstruction over the sense pad, for which the sensor has  
already made full allowance for, could suddenly be removed  
leaving the sensor with an artificially elevated reference level  
and thus become insensitive to touch. In this latter case, the  
sensor will compensate for the object's removal very quickly,  
usually in only a few seconds.  
2.1.4 DETECTION  
I
NTEGRATOR  
It is desirable to suppress detections generated by electrical  
noise or from quick brushes with an object. To accomplish  
this, the QT118H incorporates a detect integration counter  
that increments with each detection until a limit is reached,  
after which the output is activated. If no detection is sensed  
prior to the final count, the counter is reset immediately to  
zero. The required count is 4.  
The Detection Integrator can also be viewed as a 'consensus'  
filter, that requires four detections in four successive bursts  
to create an output. As the basic burst spacing is 75ms, if  
this spacing was maintained throughout all 4 counts the  
2.1.2 THRESHOLD  
CALCULATION  
Unlike the QT110 device, the internal threshold level is fixed sensor would react very slowly. In the QT118H, after an  
at one of two setting as determined by Table 1-1. These initial detection is sensed, the remaining three bursts are  
setting are fixed with respect to the internal reference level, spaced about 18ms apart, so that the slowest reaction time  
which in turn can move in accordance with the drift possible is 75+18+18+18 or 129ms and the fastest possible  
compensation mechanism..  
is 54ms, depending on where in the initial burst interval the  
contact first occurred. The response time will thus average  
92ms.  
2.1.5 FORCED  
S
ENSOR  
R
ECALIBRATION  
The QT118H has no recalibration pin; a forced recalibration  
is accomplished only when the device is powered up.  
However, the supply drain is so low it is a simple matter to  
treat the entire IC as a controllable load; simply driving the  
QT118H's Vdd pin directly from another logic gate or a  
microprocessor port (Figure 2-2) will serve as both power  
and 'forced recal'. The source resistance of most CMOS  
gates and microprocessors is low enough to provide direct  
power without any problems. Note that most 8051-based  
microcontrollers have only a weak pullup drive capability  
and will require true CMOS buffering. Any 74HC or 74AC  
series gate can directly power the QT118H, as can most  
other microcontrollers. A 0.01uF minimum bypass capacitor  
close to the device is essential; without it the device can  
Figure 2-2 Powering From a CMOS Port Pin  
PORT X.m  
0.01µF  
C MO S  
m ic rocontroller  
Vdd  
PORT X.n  
OUT  
QT118  
Vss  
lq  
4
break into high frequency oscillation, get hot, and stop  
working.  
2.2.4 HEARTBEAT™ OUTPUT  
The output has a full-time HeartBeat™ ‘health’ indicator  
superimposed on it. This operates by taking 'Out' into a  
3-state mode for 350µs once before every QT burst. This  
output state can be used to determine that the sensor is  
operating properly, or, it can be ignored using one of several  
simple methods.  
Option strap configurations are read by the QT118H only on  
powerup. Configurations can only be changed by powering  
the QT118H down and back up again; a microcontroller can  
directly alter most of the configurations and cycle power to  
put them in effect.  
Since Out is normally low, a pullup resistor will create  
positive HeartBeat pulses (Figure 2-3) when the sensor is  
not detecting an object; when detecting an object, the output  
will remain active for the duration of the detection, and no  
HeartBeat pulse will be evident.  
2.2 OUTPUT FEATURES  
The QT118H is designed for maximum flexibility and can  
accommodate most popular sensing requirements. These  
are selectable using strap options on pins OPT1 and OPT2.  
All options are shown in Table 2-1.  
If the sensor is wired to a microcontroller as shown in Figure  
2-4, the controller can reconfigure the load resistor to either  
ground or Vcc depending on the output state of the device,  
so that the pulses are evident in either state.  
2.2.1 DC MODE  
OUTPUT  
The output of the device can respond in a DC mode, where  
the output is active-high upon detection. The output will  
remain active for the duration of the detection, or until the  
Max On-Duration expires, whichever occurs first. If the latter  
occurs first, the sensor performs a full recalibration and the  
output becomes inactive until the next detection.  
Electromechanical devices will usually ignore this short  
pulse. The pulse also has too low a duty cycle to visibly  
activate LED’s. It can be filtered completely if desired, by  
adding an RC timeconstant to filter the output, or if  
interfacing directly and only to a high-impedance CMOS  
input, by doing nothing or at most adding a small non-critical  
capacitor from Out to ground (Figure 2-5).  
In this mode, two Max On-Duration timeouts are available:  
10 and 60 seconds.  
2.2.2 TOGGLE  
M
ODE  
O
UTPUT  
2.2.5 PIEZO  
A
COUSTIC  
D
RIVE  
This makes the sensor respond in an on/off mode like a flip  
flop. It is most useful for controlling power loads, for  
example in kitchen appliances, power tools, light switches,  
etc.  
A piezo drive signal is generated for use with a bare piezo  
sounder immediately after a detection is made; the tone lasts  
for a nominal 75ms to create a ‘tactile feedback’ sound.  
The sensor will drive most common bare piezo ‘beepers’  
directly using an H-bridge drive configuration for the highest  
possible sound level at all supply voltages; H-bridge drive  
effectively doubles the supply voltage across the piezo. The  
piezo is connected across pins SNS1 and SNS2. This drive  
operates at a nominal 4kHz frequency, a common resonance  
point for enclosed piezo sounders. Other frequencies can be  
obtained upon special request.  
Max On-Duration in Toggle mode is fixed at 10 seconds.  
When a timeout occurs, the sensor recalibrates but leaves  
the output state unchanged.  
2.2.3 PULSE  
MODE  
OUTPUT  
This generates a positive pulse of 75ms duration with every  
new detection. It is most useful for 2-wire operation (see  
Figure 1-2), but can also be used when bussing together  
several devices onto a common output line with the help of  
steering diodes or logic gates, in order to control a common  
load from several places.  
If desired a bare piezo sounder can be directly adhered to  
the rear of a control panel, provided that an acoustically  
resonant cavity is also incorporated to give the desired  
sound level.  
Max On-Duration is fixed at 10 seconds if in Pulse output  
mode.  
Since piezo sounders are merely high-K ceramic capacitors,  
the sounder will double as the Cs capacitor, and the piezo's  
Figure 2-3  
Getting HB pulses with a pulup resistor when not active  
Figure 2-4  
Using a micro to obtain HB pulses in either output state  
lq  
5
3.2 PIEZO SOUNDER  
Figure 2-5 Eliminating HB Pulses  
The use of a piezo sounder in place of Cs is described in the  
previous section. Piezo sounders have very high,  
uncharacterized thermal coefficients and should not be used  
if fast temperature swings are anticipated, especially at high  
gains.  
GATE OR  
MICRO INPUT  
2
3
4
7
C M OS  
OUT  
SNS2  
GAIN  
SNS1  
Co  
100pF  
5
6
3.3 OPTION STRAPPING  
OPT1  
OPT2  
The option pins Opt1 and Opt2 should never be left floating.  
If they are floated, the device will draw excess power and the  
options will not be properly read on powerup. Intentionally,  
there are no pullup resistors on these lines, since pullup  
resistors add to power drain if tied low.  
The Gain input is designed to be floated for sensing one of  
the three gain settings. It should never be connected to a  
pullup resistor or tied to anything other than Sns1 or Sns2.  
metal disc will act as the sensing electrode. Piezo transducer  
capacitances typically range from 6nF to 30nF (0.006µF to  
0.03µF) in value; at the lower end of this range an additional  
capacitor should be added to bring the total Cs across SNS1  
and SNS2 to at least 10nF, or more if Cx is large.  
Table 2-1 shows the option strap configurations available.  
3.4 POWER SUPPLY, PCB LAYOUT  
The power supply can range from 2.5 to 5.0 volts. At 3 volts  
current drain averages less than 20µA in most cases, but  
The burst acquisition process induces a small but audible  
voltage step across the piezo resonator, which occurs when  
SNS1 and SNS2 rapidly discharge residual voltage stored on  
the resonator. The resulting slight clicking sound can be  
used to provide an audible confirmation of functionality if  
desired, or, it can be suppressed by placing a non-critical 1M  
to 2M ohm bleed resistor in parallel with the resonator. The  
resistor acts to slowly discharge the resonator, preempting  
the occurrence of the harmonic-rich step (Figure 2-6).  
Figure 2-6 Damping Piezo Clicks with R  
x
+2.5 to 5  
SENSING  
ELECTRODE  
With the resistor in place, an almost inaudible clicking sound  
may still be heard, which is caused by the small charge  
buildup across the piezo device during each burst.  
1
Vdd  
2
3
4
7
5
6
OUT  
SNS1  
GAIN  
SNS2  
2.2.6 OUTPUT  
The QT118H’s `output is active high and it can source 1mA  
DRIVE  
OPT1  
OPT2  
or sink 5mA of non-inductive current.  
R
C
x
x
Care should be taken when the IC and the load are both  
powered from the same supply, and the supply is minimally  
regulated. The device derives its internal references from the  
power supply, and sensitivity shifts can occur with changes  
in Vdd, as happens when loads are switched on. This can  
induce detection ‘cycling’, whereby an object is detected, the  
load is turned on, the supply sags, the detection is no longer  
sensed, the load is turned off, the supply rises and the object  
is reacquired, ad infinitum. To prevent this occurrence, the  
output should only be lightly loaded if the device is operated  
from an unregulated supply, e.g. batteries. Detection  
‘stiction’, the opposite effect, can occur if a load is shed  
when Out is active.  
Vss  
8
can be higher if Cs is large. Large Cx values will actually  
decrease power drain. Operation can be from batteries, but  
be cautious about loads causing supply droop (see Output  
Drive, previous section).  
As battery voltage sags with use or fluctuates slowly with  
temperature, the IC will track and compensate for these  
changes automatically with only minor changes in sensitivity.  
If the power supply is shared with another electronic system,  
care should be taken to assure that the supply is free of  
digital spikes, sags, and surges which can adversely affect  
the device. The IC will track slow changes in Vdd, but it can  
be affected by rapid voltage steps.  
3 - CIRCUIT GUIDELINES  
3.1 SAMPLE CAPACITOR  
When used for most applications, the charge sampler Cs  
can be virtually any plastic film or ceramic capacitor. The  
type should be relatively stable in the anticipated  
temperature range. If fast temperature swings are expected,  
especially with higher sensitivities, more stable capacitors be  
required, for example PPS film, X7R, or NPO/C0G ceramic.  
if desired, the supply can be regulated using a conventional  
low current regulator, for example CMOS regulators that  
have nanoamp quiescent currents. Care should be taken that  
the regulator does not have a minimum load specification,  
which almost certainly will be violated by the QT118H's low  
current requirement.  
lq  
6
Since the IC operates in a burst mode, almost all  
the power is consumed during the course of each  
burst. During the time between bursts the sensor  
is quiescent.  
Figure 2-7 ESD Protection  
For proper operation a 100nF (0.1uF) ceramic  
bypass capacitor should be used between Vdd and  
Vss; the bypass cap should be placed very close  
to the device’s power pins. Without this capacitor  
the part can break into high frequency oscillation,  
get physically hot, and stop working.  
3.4.1 MEASURING  
Measuring average power consumption is  
S
UPPLY  
C
URRENT  
a
challenging task due to the burst nature of the  
device’s operation. Even a good quality RMS DMM  
will have difficulty tracking the relatively slow burst  
rate.  
shown  
(1N4150,  
BAV99  
or  
equivalent  
low-C  
The simplest method for measuring average current is to  
replace the power supply with a large value low-leakage  
electrolytic capacitor, for example 2,700µF. 'Soak' the  
capacitor by connecting it to a bench supply at the desired  
operating voltage for 24 hours to form the electrolyte and  
reduce leakage to a minimum. Connect the capacitor to the  
circuit at T=0, making sure there will be no detections during  
the measurement interval; at T=30 seconds measure the  
capacitor's voltage with a DMM. Repeat the test without a  
load to measure the capacitor's internal leakage, and  
subtract the internal leakage result from the voltage droop  
measured during the QT118H load test. Be sure the DMM is  
connected only at the end of each test, to prevent the DMM's  
impedance from contributing to the capacitor's discharge.  
high-conductance diodes) will shunt the ESD transients  
away from the part, and Re1 will current limit the rest into  
the QT118H's own internal clamp diodes. C1 should be  
around 10µF if it is to absorb positive transients from a  
human body model standpoint without rising in value by  
more than 1 volt. If desired C1 can be replaced with an  
appropriate zener diode. Directly placing semiconductor  
transient protection devices or MOV's on the sense lead is  
not advised; these devices have extremely large amounts of  
nonlinear parasitic C which will swamp the capacitance of  
the electrode.  
Re1 and Re2 should be as large as possible given the load  
value of Cx, Cf, and the diode capacitances of D1 and D2.  
Re1 and Re2 should be low enough to permit at least 6 RC  
time-constants to occur during the charge and transfer  
phases. Re1 should be about 20% of Re2. Cf is used for RFI  
suppression; see below.  
Supply drain can be calculated from the adjusted voltage  
droop using the basic charge equation:  
✁VC  
t
i =  
Re3 functions to isolate the transient from the Vdd pin;  
values of around 1K ohms are reasonable.  
where C is the large supply cap value, t is the elapsed  
measurement time in seconds, and ∆V is the adjusted  
voltage droop on C.  
As with all protection networks, it is crucial that transients be  
led away from the circuit. PCB ground layout is crucial; the  
ground connections to D1, D2, and C1 should all go back to  
the power supply ground or preferably, if available, a chassis  
ground connected to earth. The currents should not be  
allowed to traverse the area directly under the IC.  
A good approximation can be made to this method by using  
a 2,700µF cap across the circuit, and inserting a 220 ohm  
resistor in series with a current meter in the power wire.  
3.4.2 ESD, RFI  
P
ROTECTION  
If the electrode operates behind glass or insulating plastics  
thicker than 2mm, D1 and D2 can be safely deleted.  
However it is still wise to use Re1, of a value as large as can  
be tolerated. Values up to 100K and sometimes well beyond  
can usually be tolerated quite well.  
ESD protection. In cases where the electrode is placed  
behind a dielectric panel, the IC will be protected from direct  
static discharge. However even with a panel transients can  
still flow into the electrode via induction, or in extreme cases  
via dielectric breakdown. Porous materials may allow a  
spark to tunnel right through the material. Testing is required  
to reveal any problems. The device does have diode  
protection on its terminals which can absorb and protect the  
device from most induced discharges, up to 20mA; the  
usefulness of the internal clamping will depending on the  
dielectric properties, panel thickness, and rise time of the  
ESD transients.  
If the device is connected to an external circuit via a cable or  
long twisted pair, it is possible for ground-bounce to cause  
damage to the Out pin; even though the transients are led  
away from the IC itself, the connected signal or power  
ground line will act as an inductor causing a high differential  
voltage to build up on the Out wire with respect to ground. If  
this is a possibility the Out pin should have a resistance Re4  
in series with it to limit current; this resistor should be as  
large as can be tolerated by the load.  
ESD dissipation can be aided further with an added diode  
protection network as shown in Figure 2-7, in extreme cases.  
Because the charge and transfer times of the QT118H are  
relatively long, the circuit can tolerate very large values of  
Re1 and Re2, more than 100k ohms combined in most  
cases where the electrode load is small. The added diodes  
RFI Suppression. PCB layout, grounding, and the structure  
of the input circuitry have a great bearing on the success of  
a design to withstand RF fields.  
lq  
7
The circuit is remarkably immune to HF RFI provided that  
certain design rules be adhered to:  
7. The sense electrode should be kept away from other  
conductors, even ground, which can re-radiate in RF  
currents.  
1. Use SMT components.  
8. If the ESD diodes are not used, use Re1 in the electrode  
trace anyway, with a value as large as possible without  
compromising gain.  
2. Always use a ground plane under the circuit.  
3. Use a 0.1uF bypass cap very close to the supply pins.  
4. If ESD diodes are used, always use Re1, Re2, and Cf.  
Make Re1 / Re2 as large as possible without  
compromising gain (depends on Cf and Cx).  
Cf acts to shunt aside RF from entering the two diodes, thus  
preventing their conduction due to RF currents. This form of  
conduction will lead to false or erratic operation. Cf also acts  
to lower sensitivity, and in many cases Cs will need to be  
increased to compensate for this loss.  
5. Keep all ESD components close to the IC.  
6. Do not route the sense wire near other traces or wires  
lq  
8
4.1 ABSOLUTE MAXIMUM SPECIFICATIONS  
Operating temp . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . as designated by suffix  
Storage temp . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . -55OC to +125OC  
V
DD . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . -0.5 to +6.5V  
Max continuous pin current, any control or drive pin . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 20mA  
Short circuit duration to ground, any pin . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . infinite  
Short circuit duration to VDD, any pin . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . infinite  
Voltage forced onto any pin . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . -0.6V to (Vdd + 0.6) Volts  
4.2 RECOMMENDED OPERATING CONDITIONS  
DD . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . +2.5 to 5.5V  
V
Short-term supply ripple+noise . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5mV  
Long-term supply stability . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 100mV  
Cs value . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 10nF to 500nF  
Cx value . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 0 to 100pF  
4.3 AC SPECIFICATIONS Vdd = 3.0, Ta = recommended operating range  
Parameter  
Description  
Min  
Typ  
Max  
Units  
Notes  
T
RC  
PC  
Recalibration time  
550  
2
ms  
µs  
T
Charge duration  
T
T
T
PT  
BS  
BL  
Transfer duration  
Burst spacing interval  
Burst length  
2
µs  
75  
ms  
ms  
ms  
kHz  
ms  
ms  
µs  
0.5  
50  
depends heavily on Cs, Cx  
with minimal Cs  
T
R
Response time  
129  
4
F
T
P
P
Piezo drive frequency  
Piezo drive duration  
Pulse output width on Out  
Heartbeat pulse width  
75  
75  
300  
T
PO  
HB  
T
4.4 SIGNAL PROCESSING  
Description  
Min  
Typ  
Max  
Units  
Notes  
Threshold differential  
6, 12, or 24  
counts  
%
1
2
Hysteresis  
17  
4
Consensus filter length  
samples  
ms/level  
ms/level  
secs  
Positive drift compensation rate  
Negative drift compensation rate  
Post-detection recalibration timer duration (typical)  
750  
75  
4
4
10  
60  
3, 4  
Note 1: Pin options  
Note 2: Of signal threshold  
Note 3: Pin option  
Note 4: Cs, Cx dependent  
lq  
9
4.5 DC SPECIFICATIONS  
Vdd = 3.0V, Cs = 10nF, Cx = 5pF, T  
A
= recommended range, unless otherwise noted  
Parameter Description  
Min  
Typ  
Max  
Units  
Notes  
V
DD  
DD  
DDS  
Supply voltage  
2.45  
5.25  
V
µA  
V/s  
V
I
Supply current  
20  
V
Supply turn-on slope  
Low input logic level  
High input logic level  
Low output voltage  
High output voltage  
Input leakage current  
Load capacitance range  
Min shunt resistance  
Acquisition resolution  
Sensitivity range  
100  
2.2  
Required for proper startup  
OPT1, OPT2  
V
IL  
0.8  
0.6  
V
HL  
OL  
V
OPT1, OPT2  
V
V
OUT, 4mA sink  
OUT, 1mA source  
OPT1, OPT2  
V
OH  
IL  
Vdd-0.7  
0
V
I
1
µA  
pF  
✡
C
X
100  
I
X
500K  
Resistance from SNS1 to SNS2  
Note 2  
A
R
14  
28  
bits  
fF  
S
1,000  
Preliminary Data: All specifications subject to change.  
Figure 4-1 - Typical Threshold Sensitivity vs. Cx,  
High Gain, at Selected Values of Cs; Vdd = 3.0  
Figure 4-2 - Typical Threshold Sensitivity vs. Cx,  
Medium Gain, Selected Values of Cs; Vdd = 3.0  
10.00  
1.00  
0.10  
0.01  
10.00  
1.00  
0.10  
0.01  
10nF  
10nF  
20nF  
50nF  
20nF  
50nF  
100nF  
200nF  
500nF  
100nF  
200nF  
500nF  
0
10  
20  
30  
40  
0
10  
20  
30  
40  
Cx Load, pF  
Cx Load, pF  
lq  
10  
Package type: 8pin Dual-In-Line  
Millimeters  
Inches  
Max  
SYMBOL  
Min  
Max  
Notes  
Min  
Notes  
a
A
6.096  
7.62  
7.112  
8.255  
10.922  
7.62  
-
0.24  
0.3  
0.28  
0.325  
0.43  
0.3  
M
m
Q
P
9.017  
7.62  
Typical  
BSC  
0.355  
0.3  
Typical  
BSC  
0.889  
0.254  
0.355  
1.397  
2.489  
3.048  
0.381  
3.048  
-
0.035  
0.01  
0.014  
0.055  
0.098  
0.12  
0.015  
0.12  
-
-
-
-
L
0.559  
1.651  
2.591  
3.81  
-
0.022  
0.065  
0.102  
0.15  
-
L1  
F
Typical  
Typical  
R
r
S
3.556  
4.064  
7.062  
9.906  
0.381  
0.14  
0.16  
0.3  
S1  
Aa  
x
7.62  
BSC  
0.3  
BSC  
8.128  
0.203  
0.32  
0.008  
0.39  
0.015  
Y
Package type: 8pin SOIC  
Millimeters  
Max  
Inches  
Max  
SYMBOL  
Min  
Notes  
Min  
Notes  
M
W
Aa  
H
h
4.800  
5.816  
3.81  
4.979  
6.198  
3.988  
1.728  
0.762  
1.27  
0.189  
0.229  
0.15  
0.196  
0.244  
0.157  
0.068  
0.01  
0.05  
0.019  
0.04  
0.01  
0.03  
8º  
1.371  
0.101  
1.27  
0.054  
0.004  
0.050  
0.014  
0.02  
D
L
BSC  
BSC  
0.355  
0.508  
0.19  
0.483  
1.016  
0.249  
0.762  
8º  
E
e
0.007  
0.229  
0º  
ß
0.381  
0º  
Ø
lq  
11  
5 - ORDERING INFORMATION  
PART  
TEMP RANGE  
PACKAGE  
MARKING  
QT118H-D  
QT118H-S  
QT118H-IS  
0 - 70C  
0 - 70C  
PDIP  
QT1 + 118  
QT1 + 8  
SOIC-8  
SOIC-8  
-40 - 85C  
QT1 + T  
Quantum Research Group Ltd.  
www.qprox.com  
techsupport@qprox.com  
Capstan House, High Street  
Hamble, Hants SO31 4HA  
United Kingdom  
US: +1 (412) 391-7367  
UK: +44 (0)23 8045 3934  
fax: +44 (0)23 8045 3939  
QProx is a trademark of QRG Ltd.  
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