MAX16813BAUP/V+,PDF,MAX16813BAUP/V+ PDF资料,Datasheet,下载MAX16813BAUP/V+技术资料

EVALUATION KIT AVAILABLE

MAX16813B

Integrated, 4-Channel, High-Brightness

LED Driver with High-Voltage DC-DC Controller

and Battery Disconnect

General Description

Beneﬁts and Features

The MAX16813B high-efficiency, high-brightness LED

(HB LED) driver provides four integrated LED current-

sink channels. An integrated current-mode switching

controller drives a DC-DC converter that provides the

necessary voltage to multiple strings of HB LEDs. The

device accepts a wide 4.75V to 40V input voltage range

and withstands direct automotive load-dump events. The

wide input range allows powering HB LEDs for small- to

medium-sized LCD displays in automotive and general

lighting applications.

● 4-Channel Linear LED Current Sinks with Internal

MOSFETs Independently Drive Multiple LED Strings

• Full-Scale LED Current, Adjustable from 20mA

to 150mA

• Drives 1 to 4 LED Strings

• 10000:1 PWM Dimming at 200Hz

● Flexible Current-Mode Architecture Supports a Wide

Range of Applications While Minimizing Interference

• Boost or SEPIC Current-Mode DC-DC Controller

• 200kHz to 2MHz Programmable Switching

Frequency

An internal current-mode switching DC-DC controller

supports boost or SEPIC topologies and operates in an

adjustable frequency range between 200kHz and 2MHz.

An integrated spread-spectrum mode helps reduce EMI.

Current-mode control with programmable slope compen-

sation provides fast response and simplifies loop compen-

sation. An adaptive output-voltage control scheme mini-

mizes power dissipation in the LED current-sink paths.

The device has a separate p-channel drive (PGATE) pin

that is used for output undervoltage protection. Whenever

the output falls below the threshold, the external

p-MOSFET is latched off, disconnecting the input source.

Cycling the EN or the input supply is required to restart

the converter. The external p-MOSFET is off when the EN

pin is below 0.3V (typ). The shutdown current is 1µA (typ)

at an input voltage of 12V.

• External Switching-Frequency Synchronization

• Spread-Spectrum Mode

● Protection Features Enhance Fault Detection and

System Reliability

• Output-to-Ground Undervoltage Protection

• Open-Drain Fault-Indicator Output

• Open-LED and LED-Short Detection and

Protection

• Overtemperature Protection

● Adaptive Output-Voltage Optimization to Minimize

Power Dissipation

• Less than 2µA Shutdown Current

Applications

● Automotive Displays LED Backlights

The device consists of four identical linear current-sink

channels, adjustable from 20mA to 150mA with an

accuracy of ±3% using a single external resistor. Multiple

channels can be connected in parallel to achieve higher

current per LED string. The device also features a unique

pulsed dimming control through a logic input (DIM),

with minimum pulse width as low as 500ns. Protection

features include output overvoltage, open-LED detection

and protection, programmable shorted-LED detection and

protection, output undervoltage detection and protection,

and overtemperature protection. The device operates

over the -40°C to +125°C automotive temperature range.

The MAX16813B is available in 20-pin (6.5mm x 4.4mm)

TSSOP and 20-pin (4mm x 4mm) TQFN packages.

● Automotive RCL, DRL, Front Position, and Fog Lights

● LCD TV and Desktop Display LED Backlights

● Architectural, Industrial, and Ambient Lighting

Ordering Information appears at end of data sheet.

19-100144; Rev 1; 1/18

MAX16813B

Integrated, 4-Channel, High-Brightness

LED Driver with High-Voltage DC-DC Controller

and Battery Disconnect

Absolute Maximum Ratings

IN to SGND ...........................................................-0.3V to +45V

EN, PGATE to SGND...................................-0.3V to (IN + 0.3V)

PGND to SGND....................................................-0.3V to +0.3V

LEDGND to SGND...............................................-0.3V to +0.3V

OUT_ to LEDGND.................................................-0.3V to +45V

OUT_ Continuous Current..............................................±175mA

Short-Circuit Duration........................................Continuous

V

CC

Continuous Power Dissipation (T = +70°C) (Note 1)

A

20-Pin TQFN (derate 25.6mW/°C above +70°C)......2051mW

20-Pin TSSOP (derate 26.5mW/°C above +70°C)....2122mW

Operating Temperature Range......................... -40°C to +125°C

Junction Temperature......................................................+150°C

Storage Temperature Range............................ -65°C to +150°C

Lead Temperature (soldering, 10s) .................................+300°C

Soldering Temperature (reflow).......................................+260°C

V

to SGND............-0.3V to the lower of (IN + 0.3V) and +6V

CC

FLT, DIM, RSDT, OVP to SGND.............................-0.3V to +6V

CS, NDRV, RT, COMP, SETI to SGND.... -0.3V to (V + 0.3V)

NDRV Peak Current (< 100ns) .............................................±3A

NDRV Continuous Current.............................................±100mA

CC

Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings only, and functional operation of the device at these

or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to absolute maximum rating conditions for extended periods may affect

device reliability.

(Note 1)

Package Thermal Characteristics

TQFN

TSSOP

Junction-to-Ambient Thermal Resistance (θ ) .....+37.7°C/W

Junction-to-Ambient Thermal Resistance (θ ) ........+39°C/W

JA

Junction-to-Case Thermal Resistance (θ )...............+6°C/W

Junction-to-Case Thermal Resistance (θ )............+2.0°C/W

JC

Note 1: Package thermal resistances were obtained using the method described in JEDEC specification JESD51-7, using a four-layer

board. For detailed information on package thermal considerations, refer to www.maximintegrated.com/thermal-tutorial.

Package Information

For the latest package outline information and land patterns (footprints), go to www.maximintegrated.com/packages. Note that a “+”,

“#”, or “-” in the package code indicates RoHS status only. Package drawings may show a different suffix character, but the drawing

pertains to the package regardless of RoHS status.

PACKAGE TYPE

PACKAGE CODE

OUTLINE NO.

LAND PATTERN NO.

20 TSSOP-EP

20 TQFN-EP

U20E+6

T2044+3

21-0108

21-0139

90-0114

90-0037

Electrical Characteristics

(V = V

= 12V, R = 12.25kΩ, R

= 15kΩ, C

= 1µF, NDRV = COMP = OUT_ = PGATE = unconnected, V

= V

=

IN

EN

RT

SETI

VCC

RSDT

DIM

V

, V

= 0.7V, V

= V

= 0V, T = T = -40°C to +125°C, unless otherwise noted. Typical values are

CC OVP

CS

LEDGND

PGND

SGND A J

at T = +25°C.) (Note 2)

A

PARAMETER

SUPPLIES

SYMBOL

CONDITIONS

MIN

TYP

MAX

UNITS

Operating Voltage Range

V

4.75

40

V

IN

V

= 1.266V, all channels on,

= 0.5V

OVP

Supply Current

I

3.4

5.7

mA

IN

V

OUT_

Standby Supply Current

IN Undervoltage Lockout

IN UVLO Hysteresis

I

V

= 0V

1

2

µA

V

IN_Shdn

EN

IN

V

rising

3.975

4.3

170

4.625

mV

Maxim Integrated

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MAX16813B

Integrated, 4-Channel, High-Brightness

LED Driver with High-Voltage DC-DC Controller

and Battery Disconnect

Electrical Characteristics (continued)

(V = V

= 12V, R = 12.25kΩ, R

= 15kΩ, C

= 1µF, NDRV = COMP = OUT_ = PGATE = unconnected, V

= V

=

IN

EN

RT

SETI

VCC

RSDT

DIM

V

, V

= 0.7V, V

= V

= 0V, T = T = -40°C to +125°C, unless otherwise noted. Typical values are

CC OVP

CS

LEDGND

PGND

SGND A J

at T = +25°C.) (Note 2)

A

PARAMETER

REGULATOR

SYMBOL

CONDITIONS

MIN

TYP

MAX

UNITS

V

CC

6.5V < V < 10V, 1mA < I

< 50mA

< 10mA

4.75

5

5.25

500

IN

LOAD

Regulator Output Voltage

V

CC

10V < V < 40V, 1mA < I

5

IN

LOAD

Dropout Voltage

V

- V , V = 4.75V, I = 50mA

LOAD

200

100

mV

mA

IN

CC IN

Short-Circuit Current Limit

V

shorted to SGND

CC_ILIM

CC

V

Undervoltage-Lockout

CC

V

rising

4

V

CC

Threshold

V

UVLO Hysteresis

125

mV

CC

RT OSCILLATOR

Switching Frequency Range

f

Frequency dithering disabled

200

90

2000

98.5

95

kHz

%

SW

f

= 200kHz to 600kHz

= 600kHz to 2000kHz

94.5

90.5

SW

Maximum Duty Cycle

86

= 200kHz to 2000kHz, frequency dither

Oscillator Frequency Accuracy

Frequency Dither

-7.5

+7.5

-9

%

disabled

Dither enabled, f

2000kHz

= from 200kHz to

SW

f

-5

4

-7

DITH

Sync Rising Threshold

Minimum Sync Frequency

PWM COMPARATOR

V

1.1f

kHz

SW

PWM Comparator Leading-Edge

Blanking

60

90

ns

PWM-to-NDRV Propagation

Delay

Including leading-edge blanking time

SLOPE COMPENSATION

Peak Slope Compensation

Current Ramp Magnitude

Current ramp added to the CS input

(Note 3)

45

50

55

µA

CURRENT-SENSE COMPARATOR

Current-Limit Threshold

396

416

10

437

mV

ns

CS Limit Comparator to NDRV

Propagation Delay

10mV overdrive, excluding leading edge

blanking time

ERROR AMPLIFIER

OUT_ Regulation Voltage

Transconductance

No-Load Gain

1

V

g

V

= 2V

340

600

75

880

µS

dB

µA

M

COMP

(Note 4)

COMP Sink Current

V

= 2.25V, V

= 2V

160

375

800

OUT_

COMP

COMP Source Current

= 0V, V

= 1.0V

375

µA

COMP

Maxim Integrated

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MAX16813B

Integrated, 4-Channel, High-Brightness

LED Driver with High-Voltage DC-DC Controller

and Battery Disconnect

Electrical Characteristics (continued)

(V = V

= 12V, R = 12.25kΩ, R

= 15kΩ, C

= 1µF, NDRV = COMP = OUT_ = PGATE = unconnected, V

= V

=

IN

EN

RT

SETI

VCC

RSDT

DIM

V

, V

= 0.7V, V

= V

= 0V, T = T = -40°C to +125°C, unless otherwise noted. Typical values are

CC OVP

CS

LEDGND

PGND

SGND A J

at T = +25°C.) (Note 2)

A

PARAMETER

MOSFET DRIVER

SYMBOL

CONDITIONS

MIN

TYP

MAX

UNITS

I

= 100mA (nMOS)

0.9

1.1

2

Ω

SINK

NDRV On-Resistance

= 50mA (pMOS)

SOURCE

Peak Sink Current

Peak Source Current

Rise Time

V

= 5V

A

NDRV

V

= 0V

2

A

C

= 1nF

6

ns

LOAD

Fall Time

C

6

LED CURRENT SOURCE

OUT_ Current Sink Range

20

-2

150

+2

mA

%

Channel-to-Channel Matching

I

= 100mA

OUT_

R

= 30kΩ, T = +25°C

48.25

47.50

97

50

51.75

52.50

103

SETI

DIM

A

= 30kΩ, T = -40°C to +125°C

A

= 15kΩ, T = +25°C

100

150

A

OUT_ Current

mA

µA

= 15kΩ, T = -40°C to +125°C

96

104

A

= 10kΩ, T = +25°C

145.50

144

-2

154.50

156

A

= 10kΩ, T = -40°C to +125°C

A

OUT_ Leakage Current

LOGIC INPUTS and OUTPUTS

EN Input Logic-High

EN Input Logic-Low

V

= 0V, V

= 40V

+2

OUT_

2.1

V

0.4

EN Hysteresis

260

7.5

mV

µA

nA

V

= 12V

15

EN

EN Input Current

= 0.3V

100

200

EN

DIM Input Logic-High

DIM Input Logic-Low

DIM Hysteresis

2.1

-2

0.8

+2

V

250

mV

µA

ns

DIM Input Current

V

= 5V

DIM

DIM to LED Turn-On Delay

DIM to LED Turn-Off Delay

DIM rising edge to 10% rise in I

150

50

OUT_

DIM falling edge to 10% fall in I

I

Rise Time

Fall Time

10% to 90% I

90% to 10% I

200

OUT_

50

ns

V

OUT_

V

= 4.75V and I

= 5mA

SINK

0.4

+1

FLT Output Low Voltage

IN

-1

µA

V

= 5.5V

FLT Output Leakage Current

LED Short-Detection Threshold

FLT

= 2V

6.1

7

7.9

RSDT

Short-Detection Comparator

Delay

6.5

µs

Maxim Integrated

│ 4

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MAX16813B

Integrated, 4-Channel, High-Brightness

LED Driver with High-Voltage DC-DC Controller

and Battery Disconnect

Electrical Characteristics (continued)

(V = V

= 12V, R = 12.25kΩ, R

= 15kΩ, C

= 1µF, NDRV = COMP = OUT_ = PGATE = unconnected, V

= V

=

IN

EN

RT

SETI

VCC

RSDT

DIM

V

, V

= 0.7V, V

= V

= 0V, T = T = -40°C to +125°C, unless otherwise noted. Typical values are

CC OVP

CS

LEDGND

PGND

SGND A J

at T = +25°C.) (Note 2)

A

PARAMETER

RSDT Leakage Current

OVP Trip Threshold

OVP Hysteresis

SYMBOL

CONDITIONS

MIN

-600

TYP

MAX

+600

1.266

UNITS

nA

V

= 2.5V

RSDT

OVP rising

1.190

1.228

70

V

mV

nA

OVP Leakage Current

V

= 1.25V

-200

+200

OVP

OVP Undervoltage-Detection

Threshold

OVP falling, PGATE latched off

0.485

0.585

10

0.685

V

OVP Undervoltage-Detection

Delay

OVP falling

5

20

µs

Thermal-Shutdown Threshold

Thermal-Shutdown Hysteresis

PGATE DRIVER

Temperature rising

165

15

°C

PGATE On-Resistance

PGATE Soft-Start Current

PGATE Soft-Start Time

PGATE Leakage Current

R

I

= 10mA

PGATE

100

350

10

250

490

13.25

1

Ω

PGATE

Active during PGATE soft-start time

210

µA

ms

µA

6.35

V

= 12V, V

= 0V

0.01

PGATE

EN

Note 2: 100% tested at T = +25°C. All limits over temperature are guaranteed by design, not production tested.

A

Note 3: CS threshold includes slope compensation ramp magnitude.

Note 4: Gain = dV

/dV , 0.05V < V

< 0.15V.

COMP

CS

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MAX16813B

Integrated, 4-Channel, High-Brightness

LED Driver with High-Voltage DC-DC Controller

and Battery Disconnect

Typical Operating Characteristics

(V = V

= 12V, R

= 21kΩ, R

= 15kΩ, C

= 1µF, NDRV = COMP = OUT_ = PGATE = unconnected, V

= 0.7V,

IN

EN

RT

SETI

VCC

OVP

V

= V

= 0V, load = 4 strings of 7 white LEDs, T = +25°C, unless otherwise noted.)

CS

LEDGND

DIM

PGND

SGND

A

SUPPLY CURRENT

vs. SUPPLY VOLTAGE

SUPPLY CURRENT

vs. SWITCHING FREQUENCY

V

LINE REGULATION

CC

5.03

5.02

5.01

5.00

4.99

4.98

4.97

5.0

4.8

4.6

4.4

4.2

4.0

3.8

3.6

3.4

5.0

4.8

4.6

4.4

4.2

4.0

3.8

3.6

C

= 13pF

C

= 13pF

NDRV

T

= +125°C

A

T

= +125°C

= +25°C

A

T

= +25°C

= -40°C

A

T

A

= -40°C

30

5

10

15

20

V

25

(V)

35

40

5

10

15

20

V

25

(V)

30

35

40

200 400 600 800 1000 1200 1400 1600 1800 2000

(kHz)

f

IN

SW

EN THRESHOLD VOLTAGE

vs. TEMPERATURE

EN INPUT CURRENT

vs. TEMPERATURE

V

CC

LOAD REGULATION

5.02

1.6

1.5

1.4

1.3

1.2

1.1

1.0

0.9

0.8

10.0

9.5

9.0

8.5

8.0

7.5

7.0

6.5

6.0

T

A

= +125°C

V

RISING

EN

5.00

4.98

4.96

4.94

4.92

T

= +25°C

A

V

EN

FALLING

T

A

= -40°C

0

20

40

I

60

(mA)

80

100

-50 -25

0

25

50

75 100 125

-50 -25

0

25

50

75 100 125

TEMPERATURE (°C)

VCC

OUT_ LEAKAGE CURRENT

vs. TEMPERATURE

I

vs. 1/R

V

SETI

ERROR vs. TEMPERATURE

OUT(AVG)

SETI

160

140

120

100

80

0.1

0

100

10

I

= (I

I

+ I

OUT1 OUT2

+

V

= 0V

OUT(AVG)

DIM

+ I

)/4

= 40V

OUT3 OUT4

OUT_

-0.1

-0.2

-0.3

-0.4

-0.5

1

60

0.1

0.01

40

20

10

25

40

55

70

85

100

-50 -25

0

25

50

75 100 125

-50 -25

0

25

50

75 100 125

1/R

(mS)

TEMPERATURE (°C)

SETI

Maxim Integrated

│ 6

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MAX16813B

Integrated, 4-Channel, High-Brightness

LED Driver with High-Voltage DC-DC Controller

and Battery Disconnect

Typical Operating Characteristics (continued)

(V = V

= 12V, R

= 21kΩ, R

= 15kΩ, C

= 1µF, NDRV = COMP = OUT_ = PGATE = unconnected, V

= 0.7V,

IN

EN

RT

SETI

VCC

OVP

V

= V

= V = 0V, load = 4 strings of 7 white LEDs, T = +25°C, unless otherwise noted.)

CS

LEDGND

DIM

PGND

SGND A

RSDT LEAKAGE CURRENT

OVP LEAKAGE CURRENT

vs. TEMPERATURE

SWITCHING WAVEFORM AT 5kHz

(50% DUTY CYCLE) DIMMING

vs. TEMPERATURE

toc12

2.0

300

250

200

150

100

50

V

OVP

= 0.7V

1.8

1.6

1.4

1.2

1.0

0.8

0.6

0.4

0.2

0

V

LX

10V/div

0V

0A

V

RSDT

= 2.5V

I

OUT_

100mA/div

V

BOOST

10V/div

0V

-50 -25

0

25

50

75 100 125

-50 -25

0

25

50

75 100 125

40µs/div

TEMPERATURE (°C)

LED CURRENT WAVEFORM WITH

DIM ON PULSE WIDTH OF 25µs

LED CURRENT WAVEFORM WITH

DIM ON PULSE WIDTH OF 1µs

toc13

toc14

V

DIM

5V/div

DIM

5V/div

0V

0A

I

OUT_

50mA/div

OUT_

50mA/div

0A

4µs/div

200ns/div

STARTUP WAVEFORM WITH

DIM ON PULSE WIDTH < 24t

DIM ON PULSE WIDTH ≥ 24t

SW

toc15

toc16

V

IN

20V/div

IN

20V/div

0V

V

DIM

5V/div

V

DIM

5V/div

I

OUT_

100mA/div

OUT_

100mA/div

0A

0V

0A

0V

V

BOOST

20V/div

BOOST

10V/div

20ms/div

Maxim Integrated

│ 7

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MAX16813B

Integrated, 4-Channel, High-Brightness

LED Driver with High-Voltage DC-DC Controller

and Battery Disconnect

Typical Operating Characteristics (continued)

(V = V

= 12V, R

= 21kΩ, R

= 15kΩ, C

= 1µF, NDRV = COMP = OUT_ = PGATE = unconnected, V

= 0.7V,

IN

EN

RT

SETI

VCC

OVP

V

= V

= V = 0V, load = 4 strings of 7 white LEDs, T = +25°C, unless otherwise noted.)

CS

LEDGND

DIM

PGND

SGND A

STARTUP WAVEFORM OF PGATE

AND INDUCTOR CURRENT WITH

DIM CONTINUOUSLY ON

STARTUP WAVEFORM WITH

DIM CONTINUOUSLY ON

toc17

toc18

V

EN

2V/div

IN

20V/div

0V

0A

0V

V

DIM

5V/div

V

BOOST

10V/div

I

OUT_

100mA/div

V

PGATE

10V/div

V

BOOST

10V/div

I

LX

1A/div

0V

0A

20ms/div

STARTUP WAVEFORMS WITH

DELAYED DIM INPUT

OUTPUT UNDERVOLTAGE FAULT

toc20

toc19

V

IN

PGATE

0V

10V/div

0V

V

FLT

V

DIM

5V/div

V

BOOST

0V

20V/div

10V/div

V

OVP

I

BOOST

200mV/div

500mA/div

0A

1s/div

1ms/div

FUNCTIONALITY WITH DIM = 0

FOR DURATION > 38ms (TYP)

DIM LOW DETECTION PERIOD

toc21

toc22

V

IN

V

DIM

10V/div

5V/div

0V

V

DIM

BOOST

5V/div

10V/div

38ms

V

BOOST

20V/div

0V

0A

0V

0A

I

BOOST

500mA/div

100ms/div

10ms/div

Maxim Integrated

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MAX16813B

Integrated, 4-Channel, High-Brightness

LED Driver with High-Voltage DC-DC Controller

and Battery Disconnect

Pin Conﬁguration

TOP VIEW

ꢄ

15

14

13

12

11

NDRV

1

2

3

4

5

6

7

8

9

20 PGND

19 CS

PGATE

DIM

10

9

CS 16

V

18 OUT4

17 OUT3

16 LEDGND

15 OUT2

14 OUT1

13 DIM

CC

IN

SGND

PGND 17

NDRV 18

MAX16813B

EN

COMP

RT

8

RSDT

SETI

OVP

PGATE

7

19

20

6

V

EP*

5

CC

FLT

ꢄ

OVP

12 SGND

11 RSDT

1

2

3

4

EP*

SETI 10

ꢀꢅꢅꢆꢇ

ꢀꢁꢂꢃ

*EXPOSED PAD.

Pin Description

NAME

FUNCTION

TQFN TSSOP

Bias Supply Input. Connect a 4.75V to 40V supply to IN. Bypass IN to SGND with a ceramic

capacitor.

1

2

3

4

5

6

IN

EN

Enable Input. Connect EN to logic-low to shut down the device. Connect EN to logic-high or IN

for normal operation. The EN input should not be left open.

Switching Converter Compensation Input. Connect the compensation network from COMP

to SGND for current-mode control (see the Feedback Compensation section).

COMP

Oscillator Timing Resistor Connection. Connect a timing resistor (R ) from RT to SGND to

T

9

program the switching frequency according to the formula R = 7.72 x 10 /f . Apply an

T

SW

4

7

RT

AC-coupled external clock at RT to synchronize the switching frequency with an external clock.

When the oscillator is synchronized with the external clock, the spread spectrum is disabled.

Open-Drain Fault Output. FLT asserts low when an open LED, short LED, output undervoltage,

5

6

7

8

9

FLT

OVP

SETI

or thermal shutdown is detected. Connect a pullup resistor from FLT to V

.

CC

Overvoltage/Undervoltage-Threshold Adjust Input. Connect a resistor-divider from the switching

converter output to OVP and SGND. The OVP comparator reference is internally set to 1.23V.

LED Current-Adjust Input. Connect a resistor (R

) from SETI to SGND to set the current

SETI

10

through each LED string (I

), according to the formula I

= 1500/R

.

LED

SETI

LED Short Detection Threshold-Adjust Input. Connect a resistive divider from V

to RSDT and

CC

8

11

RSDT

SGND to program the LED short detection threshold. Connect RSDT directly to V

to disable

CC

LED short detection.

Maxim Integrated

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MAX16813B

Integrated, 4-Channel, High-Brightness

LED Driver with High-Voltage DC-DC Controller

and Battery Disconnect

Pin Description (continued)

NAME

FUNCTION

TQFN TSSOP

Signal Ground. SGND is the current return path connection for the low-noise analog signals.

Connect SGND, LEDGND, and PGND at a single point.

9

12

13

SGND

DIM

Digital PWM Dimming Input. Apply a PWM signal to DIM for LED dimming control. Connect DIM

10

to V

if dimming control is not used.

CC

LED String Cathode Connection 1. OUT1 is the open-drain output of the linear current sink that

controls the current through the LED string connected to OUT1. OUT1 sinks up to 150mA. If

unused, connect OUT1 to LEDGND.

11

14

OUT1

LED String Cathode Connection 2. OUT2 is the open-drain output of the linear current sink that

controls the current through the LED string connected to OUT2. OUT2 sinks up to 150mA. If

unused, connect OUT2 to LEDGND.

12

13

14

15

16

17

OUT2

LEDGND

OUT3

LED Ground. LEDGND is the return path connection for the linear current sinks. Connect

SGND, LEDGND, and PGND at a single point.

LED String Cathode Connection 3. OUT3 is the open-drain output of the linear current sink that

controls the current through the LED string connected to OUT3. OUT3 sinks up to 150mA. If

unused, connect OUT3 to LEDGND.

LED String Cathode Connection 4. OUT4 is the open-drain output of the linear current sink that

controls the current through the LED string connected to OUT4. OUT4 sinks up to 150mA. If

unused, connect OUT4 to LEDGND.

15

16

18

19

OUT4

CS

Current-Sense Input. CS is the current-sense input for the switching regulator. A sense resistor

connected from the source of the external power MOSFET to PGND sets the switching current

limit. A resistor connected between the source of the power MOSFET and CS sets the slope

compensation ramp rate (see the Slope Compensation section).

Power Ground. PGND is the switching current return path connection. Connect SGND,

LEDGND, and PGND at a single point.

17

18

19

20

—

20

1

PGND

NDRV

PGATE

Switching n-MOSFET Gate-Driver Output. Connect NDRV to the gate of the external switching

power MOSFET.

External p-MOSFET Gate connection. Connect a resistor from this pin to the external

p-MOSFET gate. Connect PGATE to PGND through a resistor (0 to 10kΩ) if not used.

2

5V Regulator Output. Bypass V

as possible to the device.

to SGND with a minimum of 1µF ceramic capacitor as close

CC

3

V

CC

Exposed Pad. Connect EP to a large-area contiguous copper ground plane for effective power

dissipation. Do not use EP as the main IC ground connection. EP must be connected to SGND.

—

EP

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MAX16813B

Integrated, 4-Channel, High-Brightness

LED Driver with High-Voltage DC-DC Controller

and Battery Disconnect

RSDT

(SHORTED-LED THRESHOLD)

FLT

PWROND

UNUSED–

STRING

DETECTOR

MAX16813B

FAULT FLAG

LOGIC

SHORT-LED

DETECTOR

OPEN-LED

DETECTOR

UV

V

CC

SHDN

TSHDN

DRIVER

NDRV

PGND

PWM

LOGIC

CLK

OUT1–

OUT4

MINIMUM

STRING

VOLTAGE

PWM

COMP

RT OSCILLATOR/RAMP

FOR SLOPE

RT

COMPENSATION

COMP

$ARRAY = 4

DRIVER

PGATE

SOFT–START

POK

g

M

0

ILIM

PWRON

R

LOGIC

IN

0.425V

0

1

UVLO

SHDN

V

CC

LEDGND

1

CLK

5V LDO

BANDGAP

MINSTR

LODIMB

MINIMUM CYCLES

BLOCK

MINSTR

_REF

THERMAL

SHUTDOWN

OVP

COMPARATOR

POK

TSHDN

UVLO

IN

0

1

DIM

INTERNAL

DPWM

EN

CS

UV

0.585V

0.185V

SS

SHDN

DETECTION

BLOCK

INPUT

BUFFER

SHDN

V

CC

V

BG

PWROND TSHDN

CS BLANKING

SSDONE

SS_REF

POK

PWRON

SS

SOFT-START

100ms

SLOPE

COMPENSATION

0.95*VBG

PWROND

SD_MIN

RAMP FROM

TSHDN

RT OSCILLATOR

SGND

OVP

SETI

Figure 1. Simplified Functional Diagram

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MAX16813B

Integrated, 4-Channel, High-Brightness

LED Driver with High-Voltage DC-DC Controller

and Battery Disconnect

Q2

L1

D1

22µH

V

IN

C1

1µF

C2

22µF

8 LEDs

PER

STRING

D2

R10

15kΩ

C7

0.047µF

D3

R1

261kΩ

Q1

R

CS

0.15Ω

R2

10kΩ

R

SCOMP

3.32kΩ

R7

1.4kΩ

NDRV

CS

OVP

PGATE

OUT1

OUT2

ENABLE INPUT

EN

IN

OUT3

OUT4

MAX16813B

V

CC

R

SETI

C3

18.2kΩ

1µF

SETI

FLT

R6

10kΩ

V

CC

DIM

COMP

R3

30.1kΩ

R

COMP

825Ω

RSDT

RT

C

COMP

2.2µF

SGND

PGND

LEDGND

RT

18.7Ω

R4

20kΩ

Figure 2. Typical Operating Circuit

The device features a constant-frequency peak current-

mode control with programmable slope compensation to

control the duty cycle of the PWM controller. The high-

current FET driver can provide up to 2A of current to the

external n-MOSFET. The DC-DC converter implemented

using the controller generates the required supply volt-

age for the LED strings from a wide input supply range.

Connect LED strings from the DC-DC converter output to

the 4-channel constant-current sink drivers that control

the current through the LED strings. A single resistor

connected from the SETI input to ground adjusts the

forward current through all 4 LED strings.

Detailed Description

The MAX16813B high-efficiency HB LED driver

integrates all the necessary features to implement a

high-performance backlight driver to power LEDs in

small- to medium-sized displays for automotive as well as

general applications. The device provides load-dump

voltage protection up to 40V in automotive applications.

The device incorporates two major blocks: a DC-DC

controller with peak-current-mode control to implement a

boost or a SEPIC-type switched-mode power supply and

a 4-channel LED driver with 20mA to 150mA constant-

current sink capability per channel. Figure 1 is the

simplified functional diagram and Figure 2 shows a

typical operating circuit.

The device features adaptive voltage control that adjusts

the converter output voltage depending on the forward

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MAX16813B

Integrated, 4-Channel, High-Brightness

LED Driver with High-Voltage DC-DC Controller

and Battery Disconnect

voltage of the LED strings. This feature minimizes the

voltage drop across the constant-current sink drivers

and reduces power dissipation in the device. The device

includes an internal 5V LDO capable of powering addi-

tional external circuitry. A logic input (EN) shuts down the

device when pulled low. When the EN pin is pulled below

0.3V (typ), the quiescent input current to the device is less

than 1µA (typ).

Current-Mode DC-DC Controller

The peak current-mode controller allows boost or SEPIC-

type converters to generate the required bias voltage

for the LED strings. The switching frequency can be

programmed over the 200kHz to 2MHz range using a

resistor connected from RT to SGND. Programmable

slope compensation is available to compensate for sub-

harmonic oscillations that occur at above 50% duty cycles

in continuous-conduction mode.

The device provides a very wide (10000:1) PWM

dimming range where a dimming pulse as narrow as

500ns is possible at a 200Hz dimming frequency. This

is made possible by a unique feature that detects short

PWM dimming input pulses and adjusts the converter

feedback accordingly.

The external n-MOSFET is turned on at the beginning

of every switching cycle. The inductor current ramps up

linearly until turned off at the peak current level set by

the feedback loop. The peak inductor current is sensed

from the voltage across the current-sense resistor (R

)

CS

connected from the source of the external n-MOSFET

to PGND. The device features leading-edge blanking

to suppress the external n-MOSFET switching noise.

A PWM comparator compares the current-sense volt-

age plus the slope-compensation signal with the output

of the transconductance error amplifier. The controller

turns off the external n-MOSFET when the voltage at

CS exceeds the error amplifier’s output voltage. This

process repeats every switching cycle to achieve peak-

current-mode control.

Advanced features include detection and string

disconnect for open-LED strings, partial or fully short-

ed strings, and unused strings. Overvoltage protection

clamps the converter output voltage to the programmed

OVP threshold in the event of an open-LED condition.

Shorted-LED string-detection and overvoltage-protection

thresholds are programmable using the RSDT and OVP

inputs, respectively. An open-drain FLT signal asserts

to indicate open-LED, shorted-LED, output undervolt-

age and overtemperature conditions. Disable individual

current sink channels by connecting the correspond-

ing OUT_ to LEDGND. In this case, FLT does not

assert indicating an open-LED condition for the disabled

channel. The device also features an overtempera-

ture protection that shuts down the controller if the die

temperature exceeds +165°C.

Error Ampliﬁer

The internal error amplifier compares an internal feedback

(FB) with an internal reference (REF) and regulates its

output to adjust the inductor current. An internal mini-

mum string detector measures the minimum-current sink

voltage with respect to SGND out of the four constant-

current sink channels. During normal operation, this mini-

mum OUT_ voltage is regulated to 1V through feedback.

The error amplifier takes 1V as the REF and the minimum

OUT_ voltage as the FB input. The amplified error at

the COMP output controls the inductor peak current to

regulate the minimum OUT_ voltage at 1V. The resulting

DC-DC converter output voltage is the highest LED string

voltage plus 1V.

There are two levels of output undervoltage protection in

the device. The first output undervoltage protection is set

at 180mV and this is enabled 43ms after power-up. If the

OVP pin is lower than 180mV after 43ms, it turns off the

converter and disconnects the p-MOSFET from the input.

The second undervoltage threshold is activated after the

soft-start period of the DC-DC converter. This is set at

585mV. If the OVP pin is below 585mV after the soft-start

period of the DC-DC converter, the converter is turned off

and the p-MOSFET disconnects the input voltage from

the LED driver. See the Startup Sequence section for

more details.

The converter stops switching when the LED strings are

turned off during PWM dimming. The error amplifier is

disconnected from the COMP output to retain the

compensation capacitor charge. This allows the converter

to settle to a steady-state level almost immediately when

the LED strings are turned on again. This unique feature

provides fast dimming response without having to use

large output capacitors.

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MAX16813B

Integrated, 4-Channel, High-Brightness

LED Driver with High-Voltage DC-DC Controller

and Battery Disconnect

If the PWM dimming on-pulse is less than or equal to 24

switching cycles, the feedback controls the voltage on

OVP so that the converter output voltage is regulated at

95% of the OVP threshold. This mode ensures that narrow

PWM dimming pulses are not affected by the response

time of the converter. During this mode, the error amplifier

remains connected to the COMP output continuously and

the DC-DC converter continues switching.

recycled. If there is no undervoltage, soft-start terminates

when the minimum current sink voltage reaches 1V (typ)

or when an internal 100ms timeout expires.

After soft-start, the device detects open LED and discon-

nects any strings with an open LED from the internal

minimum OUT_ voltage detector. The converter output

discharges to a level where the new minimum OUT_

voltage is 1V and then control is handed over to the

internal minimum OUT_ voltage detector.

Input and V

Undervoltage Lockout (UVLO)

CC

A second output undervoltage protection is enabled

100ms after the converter is enabled. A fault is detected

whenever the OVP pin falls below an internal threshold

of 585mV (typ) and the power converter is latched off

and PGATE goes high. Cycling the EN pin or the supply

is required to start up again, once the fault condition has

been removed.

The device features two undervoltage lockouts that monitor

the input voltage at IN and the output of the internal LDO

regulator at V . The device turns on after both IN and

CC

V

CC

exceed their respective UVLO thresholds. The UVLO

threshold at IN is 4.3V when IN is rising and 4.13V when

IN is falling. The UVLO threshold at V is 4V when V

CC

is rising and 3.875V when V

is falling.

CC

Oscillator Frequency/External

Synchronization

The internal oscillator frequency is programmable between

200kHz and 2MHz using a timing resistor (R ) connected

from the RT input to SGND. Use the equation below

Enable

The device is enabled using the EN logic input pin. The

EN input can handle voltages up to IN, providing flexibil-

ity in terms of control signals/supplies. To shut down the

device, drive the EN pin with a logic-low, which reduces

current consumption to 1µA (typ). Connect the EN pin to

IN if not used. EN should not be left open.

T

to calculate the value of R for the desired switching

T

frequency (f ):

SW

9

Startup Sequence

Once EN is driven high, the controller remains off until

7.72×10

R

=

T

f

SW

both IN and V

trip their rising thresholds.

CC

where f

is in Hz.

SW

Once UVLO conditions are satisfied, the driver of the

external p-MOSFET is turned on. A constant current

of 350µA (typ) flows into the PGATE pin of the device

for approximately 10ms (typ). The current flowing into

resistor R7 and capacitor C7 (see Figure 2) pulls down

the gate of the external p-MOSFET. This capacitor

controls the turn-on time of the external p-MOSFET.

Synchronize the oscillator with an external clock by

AC-coupling the external clock to the RT input. The

capacitor used for the AC-coupling should satisfy the

following relation:













9.862

-3

C

≤

− 0.144×10

µF

(

)

SYNC

R

T

After the external p-MOSFET Q2 (Figure 2) is turned

on and the 10ms timeout expires, the device detects

and then disconnects any unused current sink

channels before enabling the converter. Disable the

unused current sink channels by connecting the

corresponding OUT_ to LEDGND. This avoids asserting

the FLT output for the unused channels. The detection of

unused channels takes approximately 0.7ms (typ).

where R is in ohms.

T

The pulse width for the synchronization pulse should sat-

isfy the following relations:

t

PW

V

< 0.5

S

t

CLK













t

PW

CLK

t

0.8 −

V

+ V > 3.4

S

t

S

Once the above phase is completed, the DC-DC converter

is enabled and the soft-start is initiated. During soft-start,

the DC-DC converter output ramps up as the loop regu-

lates the voltage at the OVP pin to follow an internal ramp-

ing voltage. 33ms (typ) after the converter is enabled, the

OVP pin is monitored, and if the voltage at the OVP pin

is less than 180mV (typ), FLT is asserted low, the power

converter is turned off, the external p-MOSFET is turned

off, and they all stay off until the EN pin or the supply is

CLK

t

<

− 1.05 × t

CLK

(

)

PW

CI

t

CI

where t

is the synchronization source pulse width,

PW

t

is the synchronization clock time period, t is the

CLK

CI

programmed clock period, and V is the synchronization

pulse voltage level.

S

Maxim Integrated

│ 14

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MAX16813B

Integrated, 4-Channel, High-Brightness

LED Driver with High-Voltage DC-DC Controller

and Battery Disconnect

LED string, use two or more of the current source outputs

(OUT_) connected together to drive the string, as shown

in Figure 3.

Spread-Spectrum Mode

The device includes a unique spread-spectrum mode

(SSM) that reduces emission (EMI) at the switching

frequency and its harmonics.

LED Dimming Control

The spread spectrum uses a pseudorandom dithering

technique where the switching frequency is varied in the

range of 93% of the programmed switching frequency, to

100% of the programmed switching frequency set through

the external resistor from RT to SGND.

The device features LED brightness control using an

external PWM signal applied to DIM. A logic-high signal

on the DIM input enables all four LED current sources and

a logic-low signal disables them.

The duty cycle of the PWM signal applied to DIM also

controls the DC-DC converter’s output voltage. If the

turn-on duration of the PWM signal is less than 24 oscil-

lator clock cycles (DIM pulse width increasing), the boost

converter regulates its output based on feedback from

the OVP input. While in this mode, the converter output

voltage is regulated to 95% of the overvoltage threshold

at the OVP pin. If the turn-on duration of the PWM signal

is greater than or equal to 24 oscillator clock cycles (DIM

pulse width increasing), the converter regulates its output

so that the minimum voltage at OUT_ is 1V.

Instead of a large amount of spectral energy present at

multiples of the switching frequency, the total energy at

the fundamental and each harmonic is spread over a

wider bandwidth, reducing the energy peak.

Spread spectrum is only disabled if external synchroniza-

tion is used.

5V LDO Regulator (V

)

CC

The internal LDO regulator converts the input voltage

at IN to a 5V output voltage at V . The LDO regulator

CC

supplies up to 50mA current to provide power to internal

At power-up, if the converter has completed the soft-start

period of 100ms (typ) and the PWM signal at the DIM pin

is still low, the device regulates the output voltage based

on the feedback signal coming from the OVP pin. Once a

PWM pulse width greater than 24 oscillator clock cycles

is applied, the converter regulates its output so that the

minimum voltage at OUT_ is 1V.

control circuitry and the gate driver. Bypass V

with a minimum of 1µF ceramic capacitor as close as

possible to the device.

to SGND

CC

PWM MOSFET Driver

The NDRV output is a push-pull output with the

on-resistance of the p-MOSFET (typically 1.1Ω) and

the on-resistance of the n-MOSFET (typically 0.9Ω).

The converter output voltage is regulated to 95% of the

overvoltage threshold at the OVP pin whenever the PWM

signal at the DIM pin is forced low for a duration longer

than 38ms (typ).

NDRV swings from PGND to V

to drive an external

CC

n-MOSFET. The driver typically sources 2.0A and sinks

2.0A allowing for fast turn-on and turn-off of high gate-

charge MOSFETs.

The power dissipation in the device is mainly a function

of the average current sourced to drive the external

BOOST CONVERTER

OUTPUT

MOSFET (I

) if there are no additional loads on

VCC

V

. I

depends on the total gate charge (QG) and

CC VCC

operating frequency of the converter.

40mA TO 300mA

PER STRING

LED Current Control

The device features four identical constant-current sources

used to drive multiple HB LED strings. The current through

each one of the four channels is adjustable between

OUT1

20mA and 150mA using an external resistor (R

)

SETI

OUT2

MAX16813B

connected between SETI and SGND. Select R

the following formula:

using

SETI

OUT3

OUT4

R

= 1500/I

OUT_

SETI

where I

is the desired output current for each of

OUT_

the four channels. If more than 150mA is required in an

Figure 3. Configuration for Higher LED String Current

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MAX16813B

Integrated, 4-Channel, High-Brightness

LED Driver with High-Voltage DC-DC Controller

and Battery Disconnect

Connect the OUT_ of all channels without LED con-

nections to LEDGND before power-up to avoid OVP

triggering at startup. When an open-LED overvoltage

condition occurs, FLT is latched low. Open-LED detection

is disabled when PWM dimming pulse width is less than

24 switching clock cycles.

Fault Protections

Fault protections in the device include cycle-by-cycle

current limiting using the PWM controller, DC-DC

converter output overvoltage protection, open-LED detec-

tion, short-LED detection and protection, output under-

voltage protection, and overtemperature shutdown. An

open-drain fault flag output (FLT) goes low when an open-

LED string is detected, a shorted-LED string is detected,

an output undervoltage, or during thermal shutdown. FLT

is cleared when the fault condition is removed during

thermal shutdown and shorted LEDs. FLT is latched low

for an open-LED or output undervoltage condition, and

can be reset by cycling power or toggling the EN pin. The

thermal-shutdown threshold is +165°C and has +15°C

hysteresis.

Short-LED Detection

The device checks for shorted LEDs at each rising edge

of DIM. An LED short is detected at OUT_ if the following

condition is met:

V

OUT_

> V + 3 x V

MINSTR RSDT

where V

is the voltage at OUT_, V

is

MINSTR

OUT_

the minimum current sink voltage, and V

is the

RSDT

programmable-LED short-detection threshold set at the

RSDT input (with V less than or equal to 2.5V).

RSDT

Open-LED Management and

Overvoltage Protection

Adjust V

to a voltage less than or equal to 2.5V

RSDT

using a voltage-divider resistive network connected at

the V output, RSDT input, and SGND. Once a short is

On power-up, the device detects and disconnects any

unused current sink channels before entering the DC-DC

converter soft-start. Disable the unused current sink

channels by connecting the corresponding OUT_ to

LEDGND. This avoids asserting the FLT output for the

unused channels. After soft-start, the device detects

open LED and disconnects any strings with an open

LED from the internal minimum OUT_ voltage detector.

This keeps the DC-DC converter output voltage within

safe limits and maintains high efficiency. During normal

operation, the DC-DC converter output regulation loop

uses the minimum OUT_ voltage as the feedback input.

If any LED string is open, the voltage at the opened

CC

detected on any of the strings, the LED strings with the

short are disconnected and the FLT output flag asserts

until the device detects that the shorts are removed on

any of the following rising edges of DIM. Connect RSDT

directly to V

to always disable LED short detection.

CC

Short-LED detection is disabled when PWM dimming

pulse width is less than 24 switching clock cycles.

Applications Information

DC-DC Converter

Three different converter topologies are possible

with the DC-DC controller in the device, which has

the ground-referenced outputs necessary to use

the constant-current sink drivers. If the LED string

forward voltage is always more than the input

supply voltage range, use the boost converter

topology. If the LED string forward voltage falls within

the supply voltage range, use the buck-boost converter

topology. Buck-boost topology is implemented using

OUT_ goes to V

. The DC-DC converter output

LEDGND

voltage then increases to the overvoltage-protection

threshold set by the voltage-divider network connected

between the converter output, OVP input, and SGND. The

overvoltage-protection threshold at the DC-DC converter

output (V ) is determined using the following formula:

OVP

R1

R2





V

= 1.23 × 1+

(see Figure 2)

OVP









either

a conventional SEPIC configuration or a

coupled-inductor buck-boost configuration. The latter is

basically a flyback converter with 1:1 turns ratio. 1:1-

coupled inductors are available with tight coupling

suitable for this application. Figure 4 shows the cou-

pled-inductor buck-boost configuration. It is also pos-

sible to implement a single inductor converter using the

MAX15054 high-side FET driver.

where 1.23V (typ) is the OVP threshold. Select R1 and

R2 such that the voltage at OUT_ does not exceed

the absolute maximum rating. As soon as the DC-DC

converter output reaches the overvoltage-protection

threshold, the PWM controller is switched off setting

NDRV low. Any current sink output with V

< 300mV

OUT_

(typ) is disconnected from the minimum voltage detector.

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MAX16813B

Integrated, 4-Channel, High-Brightness

LED Driver with High-Voltage DC-DC Controller

and Battery Disconnect

The boost converter topology provides the highest effi-

ciency among the above-mentioned topologies. The

coupled-inductor topology has the advantage of not using

a coupling capacitor over the SEPIC configuration. Also,

the feedback loop compensation for SEPIC becomes

complex if the coupling capacitor is not large enough.

range, the maximum voltage needed to drive the LED

strings including the minimum 1V across the constant

LED current sink (V

), and the total output current

LED

needed to drive the LED strings (I

) as follows:

LED

I

= I

x N

LED

SRTING SRTING

where I

is the LED current per string in amperes

SRTING

Power-Circuit Design

and N

is the number of strings used.

SRTING

First select a converter topology based on the above

factors. Determine the required input supply voltage

V

IN

4.75V TO 40V

T1

(1:1)

D1

C1

UP TO 40V

C2

R1

R2

N

R

CS

SCOMP

IN NDRV

CS

OVP

EN

V

OUT1

OUT2

OUT3

OUT4

CC

C3

MAX16813B

PGATE

R

SETI

FLT

DIM

V

CC

COMP

R3

RSDT

RT

R

C

COMP

R4

SGND

PGND

LEDGND

COMP

R

T

Figure 4. Coupled-Inductor Buck-Boost Configuration

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MAX16813B

Integrated, 4-Channel, High-Brightness

LED Driver with High-Voltage DC-DC Controller

and Battery Disconnect

Calculate the maximum duty cycle (D

following equations:

) using the

IL . The recommended saturation current limit of the

P

MAX

selected inductor is 10% higher than the inductor peak

current for boost configuration. For the coupled inductor,

the saturation limit of the inductor with only one winding

For boost configuration:

(V

+ V − V

)

LED

D1

IN_MIN

conducting should be 10% higher than IL .

D

=

P

MAX

(V

+ V − V

− 0.3V)

LED

D1

DS

SEPIC Conﬁguration

For SEPIC and coupled-inductor buck-boost configurations:

(V + V

Power-circuit design for the SEPIC configuration is very

similar to a conventional design with the output voltage

referenced to the input supply voltage. For SEPIC, the

output is referenced to ground and the inductor is split into

two parts (see Figure 5 for the SEPIC configuration). One of

the inductors (L2) takes LED current as the average current

and the other (L1) takes input current as the average current.

)

D1

− 0.3V + V

LED

D

=

MAX

(V

− V

+ V

)

IN_MIN

DS

LED

D1

where V

is the forward drop of the rectifier diode in

D1

volts (approximately 0.6V), V

supply voltage in volts, and V

is the minimum input

is the drain-to-source

IN_MIN

DS

voltage of the external MOSFET in volts when it is on,

and 0.3V is the peak current-sense voltage. Initially, use

Use the following equations to calculate the average

inductor currents (IL1

, IL2 ) and peak inductor

AVG AVG

an approximate value of 0.2V for V

to calculate D

.

DS

MAX

currents (IL1 , IL2 ) in amperes:

P

Calculate a more accurate value of D

after the power

MAX

MOSFET is selected based on the maximum inductor

I

×D

1− D

×1.1

MAX

LED

IL1

=

AVG

current. Select the switching frequency (f ) depending

SW

MAX

on the space, noise, and efficiency constraints.

The factor 1.1 provides a 10% margin to account for the

converter losses:

Boost and Coupled-Inductor Conﬁgurations

In all three converter configurations, the average

inductor current varies with the input line voltage and the

maximum average current occurs at the lowest input line

voltage. For the boost converter, the average inductor

current is equal to the input current. Select the maximum

peak-to-peak ripple on the inductor current (ΔIL). The

recommended peak-to-peak ripple is 60% of the average

inductor current.

IL2

= I

AVG

LED

Assuming the peak-to-peak inductor ripple ∆IL is ±30% of

the average inductor current:

∆IL1 = IL1

x 0.3 x 2

AVG

and:

∆IL1

2

IL1 = IL1

+

P

AVG

∆IL2 = IL2

x 0.3 x 2

Use the following equations to calculate the maximum

AVG

average inductor current (IL

) and peak inductor

AVG

∆IL2

2

current (IL ) in amperes:

IL2 = IL2

+

P

AVG

I

LED

IL

=

AVG

Calculate the minimum inductance values L1

and

MIN

1− D

MAX

L2

in henries with the inductor current ripples set to

MIN

Allowing the peak-to-peak inductor ripple ∆IL to be ±30%

the maximum value as follows:

of the average inductor current:

(V − V

− 0.3V)×D

MAX

IN_MIN

DS

L1

=

∆IL = IL

x 0.3 x 2

MIN

AVG

f

× ∆IL1

SW

and

∆IL

2

(V

− V

− 0.3V)×D

IL = IL

+

IN_MIN

DS MAX

P

AVG

L2

=

MIN

f

× ∆IL2

SW

Calculate the minimum inductance value (L

) in henries

MIN

with the inductor current ripple set to the maximum value:

where 0.3V is the peak current-sense voltage. Choose

inductors that have a minimum inductance greater than

(V

− V

− 0.3V)×D

IN_MIN

DS

MAX

L

=

MIN

the calculated L1

and L2

and current rating greater

MIN

f

× ∆IL

SW

than IL1 and IL2 , respectively. The recommended

P

where 0.3V is the peak current-sense voltage. Choose

an inductor that has a minimum inductance greater

saturation current limit of the selected inductor is 10%

higher than the inductor peak current.

than the calculated L

and current rating greater than

MIN

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MAX16813B

Integrated, 4-Channel, High-Brightness

LED Driver with High-Voltage DC-DC Controller

and Battery Disconnect

For simplifying further calculations, consider L1 and L2

as a single inductor with L1 and L2 connected in parallel.

The combined inductance value and current is calculated

as follows:

Select coupling capacitor C so that the peak-to-peak

S

ripple on it is less than 2% of the minimum input supply

voltage. This ensures that the second-order effects created

by the series resonant circuit comprising L1, C , and L2 do

S

not affect the normal operation of the converter. Use the

L1

×L2

+ L2

MIN

L

=

following equation to calculate the minimum value of C :

S

MIN

L1

MIN

I

×D

MAX

× 0.02× f

SW

LED

C

≥

S

and:

V

IN_MIN

IL

AVG

= IL1

+ IL2

AVG AVG

where C is the minimum value of the coupling capacitor

S

where IL

represents the total average current through

AVG

in farads, I

is the LED current in amperes, and the

LED

both the inductors together for SEPIC configuration. Use

these values in the calculations for SEPIC configuration

in the following sections.

factor 0.02 accounts for 2% ripple.

C4

L1

D1

V

IN

C1

C2

R1

L2

Q1

R

CS

R2

R

SCOMP

NDRV

PGATE

CS

OVP

OUT1

OUT2

ENABLE

INPUT

EN

IN

OUT3

OUT4

MAX16813B

R

SETI

V

CC

V

CC

C3

R6

FLT

DIM

R3

RSDT

RT

COMP

R

COMP

RT

R4

SGND

PGND

LEDGND

C

COMP

Figure 5. SEPIC LED Driver

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MAX16813B

Integrated, 4-Channel, High-Brightness

LED Driver with High-Voltage DC-DC Controller

and Battery Disconnect

The ESR, ESL, and the bulk capacitance of the out-

put capacitor contribute to the output ripple. In most of

Slope Compensation

The device generates a current ramp for slope

compensation. This ramp current is in sync with the

switching frequency and starts from zero at the beginning

of every clock cycle and rises linearly to reach 50µA

at the end of the clock cycle. The slope-compensating

the applications, using low-ESR ceramic capacitors can

dramatically reduce the output ESR and ESL effects. To

reduce the ESL and ESR effects, connect multiple ceramic

capacitors in parallel to achieve the required bulk capaci-

tance. To minimize audible noise during PWM dimming,

the amount of ceramic capacitors on the output is usually

minimized. In this case, an additional electrolytic or tanta-

lum capacitor provides most of the bulk capacitance.

resistor, (R ), is connected between the CS input

SCOMP

and the source of the external MOSFET. This adds a

programmable ramp voltage to the CS input voltage to

provide slope compensation.

Use the following equation to calculate the value of slope

External Switching-MOSFET Selection

compensation resistance (R

):

SCOMP

The external switching MOSFET should have a voltage

rating sufficient to withstand the maximum output voltage

together with the rectifier diode drop and any possible

overshoot due to ringing caused by parasitic inductances

For boost configuration:

V

−2V

×R ×3

(

)

LED

L

IN_MIN

CS

R

=

SCOMP

× 50µA× f

× 4

and capacitances. The recommended MOSFET V

DS

MIN

SW

voltage rating is 30% higher than the sum of the maximum

output voltage and the rectifier diode drop.

For SEPIC and coupled inductor:

V

×R × 3

LED^{− V}IN_MIN

CS

(

)

The recommended continuous-drain current rating of the

MOSFET (ID), when the case temperature is at +70°C, is

greater than that calculated below:

R

=

SCOMP

L

× 50µA× f

× 4

SW

MIN

where V

and V

are in volts, R

and R

LED

are in ohms, L

IN_MIN

is in henries, and f

value of the switch current-sense resistor, (R ) can be

SCOMP CS













2

is in hertz. The

MIN

SW

ID

=

IL

×D

×1.3

RMS

AVG

MAX

CS

calculated as follows:

For boost:

The MOSFET dissipates power due to both switching

losses and conduction losses. Use the following equation

to calculate the conduction losses in the MOSFET:

D

(

× V

(

− 2V

×R ×3

CS

)

MAX

LED

4 ×L

IN_MIN

0.396 × 0.9 = I ×R

+

CS

LP

2

P

COND

= IL

x D

x R

× f

AVG

MAX DS(ON)

MN SW

where R

is the on-state drain-to-source resistance

DS(ON)

For SEPIC:

of the MOSFET. Use the following equation to calculate

the switching losses in the MOSFET:

D

(

× V

(

− V

×R

×3

)

MAX

LED

4 ×L

IN_MIN

CS

0.396 × 0.9 = I ×R

+

CS

LP

× f

2

MN SW













IL

× V

× C × f

GD SW

1

AVG

LED

P

=

×

+

SW

2

I

GOFF

where 0.396 is the minimum value of the peak current-

sense threshold. The current-sense threshold also

includes the slope-compensation component. The

minimum current-sense threshold of 0.396 is multiplied

by 0.9 to take tolerances into account.

GON

where I

and I

are the gate currents of the

GOFF

GON

MOSFET in amperes when it is turned on and turned

off, respectively. C is the gate-to-drain MOSFET

capacitance in farads.

GD

Output Capacitor Selection

Rectiﬁer Diode Selection

For all three converter topologies, the output capaci-

tor supplies the load current when the main switch is

on. The function of the output capacitor is to reduce the

converter output ripple to acceptable levels. The entire

output-voltage ripple appears across constant-current sink

outputs because the LED string voltages are stable due to

the constant current. For the device, limit the peak-to-peak

output-voltage ripple to 200mV to get stable output current.

Using a Schottky rectifier diode produces less forward drop

and puts the least burden on the MOSFET during reverse

recovery. A diode with considerable reverse-recovery time

increases the MOSFET switching loss. Select a Schottky

diode with a voltage rating 20% higher than the maximum

boost-converter output voltage and current rating greater

than that calculated in the following equation:

I

= IL

(1− D )×1.2

MAX

D

AVG

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MAX16813B

Integrated, 4-Channel, High-Brightness

LED Driver with High-Voltage DC-DC Controller

and Battery Disconnect

relationship between the RSDT voltage and the recom-

mended maximum OUT_ voltage, assuming all the active

channels are at the same voltage level.

Setting RSDT Pin Voltage

As described in the Short-LED Detection section, the

actual LED short detection threshold depends on the

RSDT pin voltage and the minimum current sink (OUT_)

voltage.

With higher OUT_ voltages, an erroneous LED short

condition can sometimes be detected when the converter

output voltage is transitioning from regulation based on

the OVP input to regulation based on the OUT_ voltages.

An optimum choice of RSDT voltage should take into

account the maximum voltage at the OUT_ pins when

the converter is regulating its output voltage based on the

OVP pin.

The plot shown here can be used when selecting the OVP

resistor divider and the RSDT voltage. It is recommended

that the RSDT voltage be chosen to be below the curve.

In general, performance is improved when the OVP resis-

tor divider is selected to set a maximum output voltage

close to the maximum LED string voltage needed in the

application.

In particular, it is recommended that the OVP resistor

divider be selected to set the output voltage of the con-

verter (when using the OVP input) so that the voltage on

the OUT_ pins does not exceed a threshold that depends

on the RSDT setting. The plot in Figure 6 shows the

MAXIMUM OUT_VOLTAGE vs. RSDT VOLTAGE

WITH V = 5V

CC

ACTIVE OUT_ PINS AT THE SAME VOLTAGE LEVEL

45

40

35

30

25

NOT

RECOMMENDED

20

15

10

5

RECOMMENDED

0

0.3

0.9

1.1

1.3

1.5

1.7

2.3

0.5

1.9

0.7

2.1

2.5

V

RSDT

(V)

Figure 6. Maximum Output Voltage vs. RSDT Voltage

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MAX16813B

Integrated, 4-Channel, High-Brightness

LED Driver with High-Voltage DC-DC Controller

and Battery Disconnect

33ms, the p-MOSFET has to sustain the highest input

voltage and the programmed current limit.

External Disconnect MOSFET Selection

An external p-MOSFET can be used to disconnect the

boost output from the battery in the event of an output

overload or short condition. In the case of the SEPIC or

buck-boost, this protection is not necessary and in those

cases there is no need for the p-MOSFET. Connect the

PGATE pin to ground in the case of the SEPIC and buck-

boost. If it is necessary to have an output short protection

for the boost even at power-up, then the current through

the p-MOSFET (Figure 7) has to be sensed. Once the

current-sense voltage exceeds a certain threshold, it

should limit the input current to the programmed threshold.

This threshold should be set at a sufficiently high level so

that it never trips at startup or under normal operating con-

ditions. Check the safe operating area of the p-MOSFET

so that the current-limit trip threshold and the voltage on

the MOSFET do not exceed the limits of the SOA curve of

the p-MOSFET at the highest operating temperature. The

current-limit protection circuit is active for 33ms before

the short trip threshold is triggered in the device, discon-

necting the p-MOSFET from the input source. During the

Overvoltage Protection

The minimum overvoltage-protection threshold at the

DC-DC converter output (V

following formula:

) is determined using the

OVP

V

= (1.19 - OVP Hysteresis) x (1 + R1/R2)

OVPmin

volts (see Figure 2) where 1.19V is the minimum over-

voltage threshold and OVP hysteresis is 70mV. Set this

minimum overvoltage threshold so that at 92% of this

threshold the circuit can still regulate the current in the

LED string when the forward-voltage drop on all the LEDs

in the LED string are at the maximum. Use the following

formula to calculate the minimum overvoltage-threshold

set point:

V

+ 1 = 0.92 x V

OVPmin

LEDmax

where V

is the maximum voltage drop that can

LEDmax

occur on LED string.

Q2

R11

L1

D1

V

IN

TO LED STRINGS

C1

D2

R12

D3

C7

Q1

C8

R10

R

CS

Q3

R7

R

SCOMP

PGATE NDRV

CS

IN

MAX16813B

Figure 7. External Disconnect MOSFET

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MAX16813B

Integrated, 4-Channel, High-Brightness

LED Driver with High-Voltage DC-DC Controller

and Battery Disconnect

where f is in hertz, V

is in volts, I

is in amperes,

LED

Feedback Compensation

P1

LED

and C

is in farads. Compensation components

OUT

During normal operation, the feedback control loop

regulates the minimum OUT_ voltage to 1V when LED

string currents are enabled during PWM dimming. When

LED currents are off during PWM dimming, the control

loop turns off the converter and stores the steady-state

condition in the form of capacitor voltages, mainly the

output filter capacitor voltage and compensation capacitor

voltage. When the PWM dimming pulses are less than 24

switching clock cycles, the feedback loop regulates the

converter output voltage to 95% of the OVP threshold.

(R

and C ) perform two functions. C

COMP COMP

COMP

introduces a low-frequency pole that presents a -20dB/

decade slope to the loop gain. R flattens the gain

COMP

of the error amplifier for frequencies above the zero

formed by R and C . For compensation, this

COMP

zero is placed at the output pole frequency (f ) so that it

provides a -20dB/decade slope for frequencies above f

to the combined modulator and compensator response.

P1

The value of R needed to fix the total loop gain at

COMP

f

so that the total loop gain crosses 0dB with -20dB/

P1,

The worst-case condition for the feedback loop is when

the LED driver is in normal mode regulating the minimum

OUT_ voltage to 1V. The switching converter small-signal

transfer function has a right-half plane (RHP) zero for

boost configuration if the inductor current is in continuous-

conduction mode. The RHP zero adds a 20dB/decade

gain together with a 90° phase lag, which is difficult to

compensate.

decade slope at 1/5 the R

as follows:

zero frequency, is calculated

HP

For boost configuration:

f

×R

×I

CS LED

ZRHP

R

=

COMP

5 × f × GM

× V

×(1− D

)

MAX

P1

COMP

LED

For SEPIC and coupled-inductor buck-boost configurations:

The worst-case RHP zero frequency (f

as follows:

) is calculated

f

×R

×I

× V

×D

ZRHP

CS LED MAX

R

=

COMP

5 × f × GM

×(1− D

)

MAX

P1

COMP

LED

For boost configuration:

where R

is the compensation resistor in ohms,

COMP

2

)

V

(1− D

MAX

f

and f are in hertz, R is the switch current-sense

LED

ZRHP

P1 CS

f

=

ZRHP

2π ×L ×I

resistor in ohms, and GM

of the error amplifier (600μS).

is the transconductance

COMP

LED

For SEPIC and coupled-inductor buck-boost configurations:

2

The value of C is calculated as follows:

COMP

V

(1− D

)

MAX

×D

LED

f

=

ZRHP

1

2π ×L ×I

C

=

LED

MAX

COMP

2π ×R

× f

Z1

COMP

where f

is in hertz, V

is in volts, L is the

ZRHP

LED

where f

is the compensation zero placed at 1/5 of

the crossover frequency that is, in turn, set at 1/5 of the

. If the output capacitors do not have low ESR, the

ESR zero frequency may fall within the 0dB crossover

frequency. An additional pole may be required to cancel

out this pole placed at the same frequency. This is

usually implemented by connecting a capacitor in parallel

Z1

inductance value of L1 in henries, and I

is in amperes.

LED

A simple way to avoid this zero is to roll off the loop gain

to 0dB at a frequency less than 1/5 of the RHP zero

frequency with a -20dB/decade slope.

f

ZRHP

The switching converter small-signal transfer function

also has an output pole. The effective output impedance,

together with the output filter capacitance, determines the

with C

and R

. Figure 5 shows the SEPIC

COMP

output pole frequency (f ) that is calculated as follows:

P1

configuration and Figure 4 shows the coupled-inductor

buck-boost configuration.

For boost configuration:

I

LED

f

=

Design Veriﬁcation

The following criteria must be satisfied before the design

can go into production:

P1

2× π × V

× C

OUT

LED

For SEPIC and coupled-inductor buck-boost configurations:

1) The chosen inductor must not saturate at the lowest

input line voltage and the maximum output current

condition. The inductor must not saturate at the high-

est operating case temperature. Adequate margin

should be provided.

I

×D

LED

MAX

× C

OUT

f

=

P1

2× π × V

LED

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MAX16813B

Integrated, 4-Channel, High-Brightness

LED Driver with High-Voltage DC-DC Controller

and Battery Disconnect

2) Verify that the slope compensation is adequate.

Inadequate slope compensation can cause subhar-

monic oscillation. For more information on select-

ing the proper slope-compensation resistor, see the

SSlope Compensation section.

on the boost will change. The boost output voltage drops

when there is a transition from low dim to normal dim

made. If the closed-loop phase margin is less than 45°,

the output voltage might ring when the transition from LO

dim to normal dim occurs. This can cause flicker of the

LEDs and this flicker needs to be prevented by increasing

the phase margin. If the flicker is still present even when

the phase margin exceeds 60°, it may be necessary to

increase the output capacitor.

3) At the lowest input line voltage and the maximum

power condition, the signal on the CS pin should be

close to the current-limit voltage on the CS pin.

4) Select Schottky diodes, MOSFETs, and resistors that

meet the power and voltage ratings.

TEST

5) Select input and output capacitors that meet ripple-

voltage and ripple-current requirements.

RINJ

REF

6) Set the overvoltage at the appropriate point.

7) After the compensation values are designed, verify the

design by measuring the loop stability.

1N4148W

OUTPUT

Loop-Stability Veriﬁcation

C2

To verify the loop stability, it is a good idea to use a loop

analyzer to study the closed-loop gain and phase with

frequency. To check the closed-loop gain, connect the

test and reference probes of the analyzer, as shown in

Figure 8.

R1

R2

LED

STRINGS

Check the voltages on the OUT_ pins with dimming at

100% duty cycle. Then insert a diode and the injection

resistor in the string where the OUT_ voltage is closest

to 1V. The added diode in series with the LED string

keeps the string where the injection resistor is added as

the string that controls the output voltage. Use an injec-

tion transformer to insert the injection voltage from test to

ref. The loop analyzer can plot the gain and phase of the

TO OVP

TO OUT1

TO OUT2

TO OUT3

TO OUT4

closed loop where the loop gain is T /R . The cross-

JW JW

Figure 8. Loop Analyzer Connection to MAX16813B Circuit

over frequency occurs at the frequency where the gain is

0db. The phase margin at that frequency should exceed

45° for guaranteed stable operation. The optimum phase

margin should exceed 60°. An example of the closed-loop

gain and phase margin on a MAX16813B boost is shown

in Figure 9. This measurement was done on the typical

application shown in Figure 2 at an input voltage of 12V.

100

80

60

40

20

0

200

150

100

50

TR1: MAG (GAIN)

TR2: PHASE (GAIN)

The crossover frequency (f ) in the design is 12kHz and

C

0

the phase margin is 74°. It is important to verify the loop

stability and phase margin before the design goes into

production. The typical crossover frequency should be in

-20

-40

-50

-100

-150

-200

the range of f

/10 > f > f

/20 where f is the cross-

SW C

-60

SW

C

over frequency. The phase margin should exceed 60° if

possible. It is also important to check the performance of

the design at the transition point from low dim to high dim

and vice versa. When the device is switching over from

low DIM mode to normal DIM mode, the output voltage

-80

-100

2

10

3

4

5

10

f/Hz

Figure 9. Closed-Loop Gain and Phase Margin

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MAX16813B

Integrated, 4-Channel, High-Brightness

LED Driver with High-Voltage DC-DC Controller

and Battery Disconnect

rectifier diode, and current-sense resistor). Connect

PGND to the power ground plane as close as possible

to PGND. Connect all other ground connections to the

Analog Dimming Using External

Control Voltage

Connect a resistor (R ) to the SETI input as shown in

SETI2

power ground plane using vias close to the terminals.

Figure 10 for controlling the LED string current using an

external control voltage. The device applies a fixed 1.23V

bandgap reference voltage at SETI and measures the

current through SETI. This measured current multiplied by

a factor of 1220 is the current through each one of the four

constant-current sink channels. Adjust the current through

SETI to get analog dimming functionality by connecting

the external control voltage to SETI through the resistor

3) There are two loops in the power circuit that carry

high-frequency switching currents. One loop is when

the MOSFET is on (from the input filter capacitor

positive terminal, through the inductor, the internal

MOSFET, and the current-sense resistor, to the input

capacitor negative terminal). The other loop is when

the MOSFET is off (from the input capacitor positive

terminal, through the inductor, the rectifier diode,

output filter capacitor, to the input capacitor nega-

tive terminal). Analyze these two loops and make the

loop areas as small as possible. Wherever possible,

have a return path on the power ground plane for the

switching currents on the top-layer copper traces, or

through power components. This reduces the loop

area considerably and provides a low-inductance path

for the switching currents. Reducing the loop area also

reduces radiation during switching.

(R

). The resulting change in the LED current with

SETI2

the control voltage is linear and inversely proportional.

The LED current control range remains between 20mA

to 150mA.

Use the following equation to calculate the LED current

set by the control voltage applied:

1.23 − V

(

)

×1220

1500

C

I

=

+

OUT

R

SETI2

SETI

PCB Layout Considerations

4) Connect the power ground plane for the constant-

current LED driver portion of the circuit to LEDGND

as close as possible to the device. Connect SGND to

PGND at the same point.

LED driver circuits based on the MAX16813B device use

a high-frequency switching converter to generate the

voltage for LED strings. Take proper care while laying

out the circuit to ensure proper operation. The switching-

converter part of the circuit has nodes with very fast

voltage changes that could lead to undesirable effects

on the sensitive parts of the circuit. Follow the guidelines

below to reduce noise as much as possible:

MAX16813B

1) Connect the bypass capacitor on V

as close as

CC

R

SETI2

SETI

possible to the device and connect the capacitor

ground to the analog ground plane using vias close to

the capacitor terminal. Connect SGND of the device to

the analog ground plane using a via close to SGND.

Lay the analog ground plane on the inner layer, prefer-

ably next to the top layer. Use the analog ground plane

to cover the entire area under critical signal compo-

nents for the power converter.

1.23V

R

V

C

SETI

2) Have a power ground plane for the switching-converter

power circuit under the power components (input filter

capacitor, output filter capacitor, inductor, MOSFET,

Figure 10. Analog Dimming with External Control Voltage

Maxim Integrated

│ 25

www.maximintegrated.com

MAX16813B

Integrated, 4-Channel, High-Brightness

LED Driver with High-Voltage DC-DC Controller

and Battery Disconnect

Ordering Information

Chip Information

PROCESS: CMOS

PART

TEMP RANGE

PIN-PACKAGE

MAX16813BATP/V+

MAX16813BAUP/V+

-40°C to +125°C 20 TQFN-EP*

-40°C to +125°C 20 TSSOP-EP*

/V denotes an automotive qualified part.

+Denotes a lead(Pb)-free/RoHS-compliant package.

*EP = Exposed pad.

Maxim Integrated

│ 26

www.maximintegrated.com

MAX16813B

Integrated, 4-Channel, High-Brightness

LED Driver with High-Voltage DC-DC Controller

and Battery Disconnect

Revision History

REVISION REVISION

PAGES

DESCRIPTION

CHANGED

NUMBER

DATE

0

1

8/17

Initial release

Removed future product status from MAX16813BAUP/V+ in Ordering Information

—

1/18

26

For pricing, delivery, and ordering information, please contact Maxim Direct at 1-888-629-4642, or visit Maxim Integrated’s website at www.maximintegrated.com.

Maxim Integrated cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim Integrated product. No circuit patent licenses

are implied. Maxim Integrated reserves the right to change the circuitry and speciﬁcations without notice at any time. The parametric values (min and max limits)

shown in the Electrical Characteristics table are guaranteed. Other parametric values quoted in this data sheet are provided for guidance.

©

Maxim Integrated and the Maxim Integrated logo are trademarks of Maxim Integrated Products, Inc.

2018 Maxim Integrated Products, Inc.

│ 27

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