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QT320

型号：

QT320

描述：

2通道PROGAMMABLE先进的传感器IC[ 2 CHANNEL PROGAMMABLE ADVANCED SENSOR IC ]

品牌：

QUANTUM[ QUANTUM RESEARCH GROUP ]

页数：

18 页

PDF大小：

953 K

下载PDF手册
LQ  
QPROX™ QT320  
2-CHANNEL  
P
ROGAMMABLE  
A
DVANCED  
SENSOR IC  
ꢀ Two channel digital advanced capacitive sensor IC  
Projects two ‘touch buttons’ through any dielectric  
ꢀ
ꢀ Cloning for user-defined sensing behavior  
ꢀ 100% autocal - no adjustments required  
ꢀ
ꢀ
ꢀ
Only one external capacitor per channel  
User-defined drift compensation, threshold levels  
Variable gain via Cs capacitor change  
ꢀ Selectable output polarities  
ꢀ
ꢀ
ꢀ
Toggle mode / normal mode outputs  
HeartBeat™ health indicator on outputs (can be disabled)  
1.8 ~ 5V supply, 60µA  
APPLICATIONS  
ꢀ Light switches ꢀ Appliance control  
ꢀ Industrial panels ꢀ Security systems  
ꢀ Access systems  
ꢀ Pointing devices  
ꢀ Computer peripherals  
ꢀ Entertainment devices  
The QT320 charge-transfer (“QT’”) touch sensor chip is a self-contained digital IC capable of detecting near-proximity or  
touch on two sensing channels. It will project sense fields through almost any dielectric, like glass, plastic, stone, ceramic,  
and most kinds of wood. It can also turn small metal-bearing objects into intrinsic sensors, making them respond to proximity  
or touch. This capability coupled with its ability to self calibrate continuously can lead to entirely new product concepts.  
It is designed specifically for human interfaces, like control panels, appliances, security systems, lighting controls, or  
anywhere a mechanical switch or button may be found; it may also be used for some material sensing and control  
applications provided that the presence duration of objects does not exceed the recalibration time-out interval.  
The IC requires only a common inexpensive capacitor per channel in order to function.  
Power consumption and speed can be traded off depending on the application; drain can be as low as 60µA, allowing  
operation from batteries.  
The IC’s RISC core employs signal processing techniques pioneered by Quantum; these are specifically designed to make  
the device survive real-world challenges, such as ‘stuck sensor’ conditions and signal drift. Even sensitivity is digitally  
determined. All key operating parameters can be set by the designer via the onboard eeprom which can be configured to alter  
sensitivity, drift compensation rate, max on-duration, output polarity, and toggle mode independently on each channel.  
No external switches, opamps, or other analog components aside from Cs are usually required.  
The Quantum-pioneered HeartBeat™ signal is also included, allowing a host controller to monitor the health of the QT320  
continuously if desired; this feature can be disabled via the cloning process.  
By using the charge transfer principle, the IC delivers a level of performance clearly superior to older technologies in a highly  
cost-effective package.  
AVAILABLE OPTIONS  
TA  
00C to +700C  
-400C to +850C  
SOIC  
-
8-PIN DIP  
QT320-D  
-
QT320-IS  
LQ  
Copyright © 2002 QRG Ltd  
QT320/R1.03 08/02  
which requires several consecutive confirmations of a  
detection before an output is activated.  
Table 1-1 Pin Descriptions  
Pin  
1
Name  
OUT1  
S2B  
Function  
Detection output, Ch. 1  
Sense Ch 2 pin B  
The two channels of sensing operate in a completely  
independent fashion. A unique cloning process allows the  
internal eeprom of the device to be programmed for each  
channel, to permit unique combinations of sensing and  
processing functions for each.  
2
3
S1A  
VSS  
Sense Ch 1 pin A  
4
Negative supply (ground)  
Sense Ch 1 pin B  
Sense Ch 2 pin A  
5
S1B  
S2A  
The two sensing channels operate in interleaved  
time-sequence and thus cannot interfere with each other.  
6
7
OUT2  
VDD  
Detection output, Ch. 2  
Positive supply  
8
Alternate Pin Functions for Cloning  
3
6
7
SCK  
SDO  
SDI  
Serial clone data clock  
Serial clone data out  
Serial clone data in  
1 - OVERVIEW  
The QT320 is a 2 channel digital burst mode charge-transfer  
(QT) sensor designed specifically for touch controls; it  
includes all hardware and signal processing functions  
necessary to provide stable sensing under a wide variety of  
changing conditions. Only two low-cost, non-critical capacitors  
are required for operation.  
A unique aspect of the QT320 is the ability of the designer to  
‘clone’ a wide range of user-defined setups into the part’s  
eeprom during development and in production. Cloned setups  
can dramatically alter the behavior of each channel,  
Figure 1-1 Basic QT320 circuit  
independently. For production, the parts can be cloned  
in-circuit or can be procured from Quantum pre-cloned.  
1.2 ELECTRODE DRIVE  
1.2.1 SWITCHING  
O
The IC implements two channels of direct-to-digital  
PERATION  
Figure 1-1 shows the basic QT320 circuit using the device,  
with a conventional output drive and power supply  
connections.  
capacitance acquisition using the charge-transfer method, in  
a process that is better understood as a capacitance-  
to-digital converter (CDC). The QT switches and charge  
measurement functions are all internal to the IC (Figure 1-2).  
1.1 BASIC OPERATION  
The QT320 employs bursts of variable-length charge-transfer  
cycles to acquire its signal. Burst mode permits power  
consumption in the microamp range, dramatically reduces RF  
emissions, lowers susceptibility to EMI, and yet permits  
excellent response time. Internally the signals are digitally  
processed to reject impulse noise using a 'consensus' filter  
The CDC treats sampling capacitor Cs as a floating store of  
accumulated charge which is switched between the sense  
pins; as a result, the sense electrode can be connected to  
either pin with no performance difference. In both cases the  
rule Cs >> Cx must be observed for proper operation. The  
polarity of the charge build-up across Cs during a burst is the  
same in either case. Typical values of Cs range from 2nF to  
100nF for touch operation.  
Larger values of Cx cause charge to be transferred into Cs  
more rapidly, reducing available resolution and resulting in  
lower gain. Conversely, larger values of Cs reduce the rise of  
differential voltage across it, increasing available resolution  
and raising gain. The value of Cs can thus be increased to  
allow larger values of Cx to be tolerated (Figures 5-1 to 5-4).  
As Cx increases, the length of the burst decreases resulting in  
lower signal numbers.  
It is possible to connect separate Cx and Cx’ loads to Sa and  
Sb simultaneously, although the result is no different than if  
the loads were connected together at Sa (or Sb). It is  
important to limit the amount of stray Cx capacitance on both  
terminals, especially if the load Cx is already large. This can  
be accomplished by minimising trace lengths and widths.  
Figure 1-2 Internal Switching  
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QT320/R1.03 08/02  
1.2.2 CONNECTION TO  
E
LECTRODES  
1.3.2 KIRCHOFF  
’
S
C
The PCB traces, wiring, and any components associated with Like all capacitance sensors, the QT320 relies on Kirchoff’s  
URRENT  
L
AW  
or in contact with Sa and Sb of either channel will become  
touch sensitive and should be treated with caution to limit the  
touch area to the desired location.  
Current Law (Figure 1-4) to detect the change in capacitance  
of the electrode. This law as applied to capacitive sensing  
requires that the sensor’s field current must complete a loop,  
returning back to its source in order for capacitance to be  
sensed. Although most designers relate to Kirchoff’s law with  
regard to hardwired circuits, it applies equally to capacitive  
field flows. By implication it requires that the signal ground  
and the target object must both be coupled together in some  
manner in order for the sensor to operate properly. Note that  
there is no need to provide an actual hardwired ground  
connection; capacitive coupling to ground (Cx1) often is  
sufficient, even if the coupling might seem very tenuous. For  
example, powering the sensor via an isolated transformer will  
almost always provide ample ground coupling, since there is  
plenty of capacitance between the primary and secondary  
windings via the transformer core and from there to the power  
wiring itself directly to 'local earth'. Even when battery  
powered, just the physical size of the PCB and the object into  
which the electronics is embedded is often enough to couple  
enough back to local earth.  
Multiple touch electrodes can be connected to one sensing  
channel, for example to create a control button on both sides  
of an object, however it is impossible for the sensor to  
distinguish between the two connected touch areas.  
The implications of Kirchoff’s law can be most visibly  
demonstrated by observing the E3B eval board’s sensitivity  
change between laying the board on a table versus holding  
the board in your hand by it’s batteries. The effect can also be  
observed by holding the board only by one electrode, letting it  
recalibrate, then touching the battery end; the board will work  
quite well in this mode.  
Figure 1-3 Mesh Electrode Geometry  
1.2.3 BURST  
M
ODE  
O
PERATION  
1.3.3 VIRTUAL  
C
APACITIVE  
G
ROUNDS  
The acquisition process occurs in bursts (Figure 1-7) of  
variable length, in accordance with the single-slope CDC  
method. The burst length depends on the values of Cs and  
Cx. Longer burst lengths result in higher gains and more  
sensitivity for a given threshold setting, but consume more  
average power and are slower.  
When detecting human contact (e.g. a fingertip), grounding of  
the person is never required, nor is it necessary to touch an  
exposed metal electrode. The human body naturally has  
several hundred picofarads of ‘free space’ capacitance to the  
local environment (Cx3 in Figure 1-4), which is more than two  
orders of magnitude greater than that required to create a  
return path to the QT320 via earth. The QT320's PCB  
however can be physically quite small, so there may be little  
‘free space’ coupling (Cx1 in Figure 1-4) between it and the  
environment to complete the return path. If the QT320 circuit  
ground cannot be grounded via the supply connections, then  
Burst mode operation acts to lower average power while  
providing a great deal of signal averaging inherent in the CDC  
process, making the signal acquisition process more robust.  
The QT method is a very low impedance method of sensing  
as it loads Cx directly into a very large capacitor (Cs). This  
results in very low levels of RF susceptibility.  
1.3 ELECTRODE DESIGN  
1.3.1 ELECTRODE  
G
EOMETRY AND  
S
IZE  
There is no restriction on the shape of the electrodes; in most  
cases common sense and a little experimentation can result  
in a good electrode design. The QT320 will operate equally  
well with long, thin electrodes as with round or square ones;  
even random shapes are acceptable. The electrode can also  
be a 3-dimensional surface or object. Sensitivity is related to  
electrode surface area, orientation with respect to the object  
being sensed, object composition, and the ground coupling  
quality of both the sensor circuit and the sensed object.  
Smaller electrodes will have less sensitivity than large ones.  
If a relatively large electrode surfaces are desired, and if tests  
show that an electrode has a high Cx capacitance that  
reduces the sensitivity or prevents proper operation, the  
electrode can be made into a mesh (Figure 1-3) which will  
have a lower Cx than a solid electrode area.  
Figure 1-4 Kirchoff’s Current Law  
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QT320/R1.03 08/02  
1.4 SENSITIVITY ADJUSTMENTS  
There are three variables which influence sensitivity  
independently for each channel:  
1. Cs (sampling capacitor)  
2. Cx (unknown capacitance)  
3. Signal threshold value  
Sense  
wire  
Sense  
wire  
There is also a sensitivity dependence of the whole device on  
Vdd. Cs and Cx effects are covered in Section 1.2.1.  
The threshold setting can be adjusted independently for each  
channel from 1 to 16 counts of signal swing (Section 2.2).  
Note that sensitivity is also a function of other things like  
electrode size, shape, and orientation, the composition and  
aspect of the object to be sensed, the thickness and  
Unshielded  
Electrode  
Shielded  
Electrode  
composition of any overlaying panel material, and the degree  
of mutual coupling of the sensor circuit and the object (usually  
via the local environment, or an actual galvanic connection).  
Figure 1-5 Field Shielding & Shaping  
It is advisable to set the sensitivity to the approximate desired  
result by changing Cx and Cs first using a signal threshold  
fixed at 10. Use the threshold value thereafter to fine-tune  
sensitivity.  
a ‘virtual capacitive ground’ may be required to increase  
return coupling.  
A ‘virtual capacitive ground’ can be created by connecting the  
QT320’s own circuit ground to:  
1.4.1 INCREASING  
S
ENSITIVITY  
In some cases it may be desirable to greatly increase  
sensitivity, for example when using the sensor with very thick  
panels having a low dielectric constant, or when sensing low  
capacitance objects.  
(1) A nearby piece of metal or metallized housing;  
(2) A floating conductive ground plane;  
(3) A fastener to a supporting structure;  
(4) A larger electronic device (to which its output might be  
connected anyway).  
Sensitivity can be increased by using a bigger electrode,  
reducing panel thickness, or altering panel composition.  
Increasing electrode size can have diminishing returns, as  
high values of Cx load will also reduce sensor gain (Figures  
5-1 to 5-4). The value of Cs also has a dramatic effect on  
sensitivity, and this can be increased in value up to a limit.  
Because the QT320 operates at a relatively low frequency,  
about 500kHz, even long inductive wiring back to ground will  
usually work fine.  
Free-floating ground planes such as metal foils should  
maximise exposed surface area in a flat plane if possible. A  
square of metal foil will have little effect if it is rolled up or  
crumpled into a ball. Virtual ground planes are more effective  
and can be made smaller if they are physically bonded to  
other surfaces, for example a wall or floor.  
Increasing electrode surface area will not substantially  
increase sensitivity if its area is already larger than the object  
to be detected. The panel or other intervening material can be  
made thinner, but again there are diminishing rewards for  
doing so. Panel material can also be changed to one having a  
higher dielectric constant, which will help propagate the field.  
Locally adding some conductive material to the panel  
(conductive materials essentially have an infinite dielectric  
constant) will also help; for example, adding carbon or metal  
fibers to a plastic panel will greatly increase frontal field  
1.3.4 FIELD  
S
HIELDING AND  
S
HAPING  
The electrode can be prevented from sensing in undesired  
directions with the assistance of metal shielding connected to  
circuit ground (Figure 1-5). For example, on flat surfaces, the  
field can spread laterally and create a larger touch area than  
desired. To stop field spreading, it is only necessary to  
surround the touch electrode on all sides with a ring of metal  
connected to circuit ground; the ring can be on the same or  
opposite side from the electrode. The ring will kill field  
spreading from that point outwards.  
If one side of the panel to which the electrode is fixed has  
moving traffic near it, these objects can cause inadvertent  
detections. This is called ‘walk-by’ and is caused by the fact  
that the fields radiate from either surface of the electrode  
equally well. Again, shielding in the form of a metal sheet or  
foil connected to circuit ground will prevent walk-by; putting a  
small air gap between the grounded shield and the electrode  
will keep the value of Cx lower and is encouraged. In the case  
of the QT320, sensitivity can be high enough (depending on  
Cx and Cs) that 'walk-by' signals are a concern; if this is a  
problem, then some form of rear shielding may be required.  
Figure 1-6 Circuit with Csx gain equalization capacitor  
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QT320/R1.03 08/02  
Figure 1-7 Burst lengths without Csx installed  
(observed using a 750K resistor in series with probe)  
Figure 1-8 Burst lengths with Csx installed  
(observed using a 750K resistor in series with probe)  
strength, even if the fiber density is too low to make the  
plastic electrically conductive.  
capacitors. This can be useful in some designs where one  
more sensitive channel is desired, but if equal sensitivity is  
required a few basic rules should be followed:  
1.4.2 DECREASING  
S
ENSITIVITY  
1. Use a symmetrical PCB layout for both channels: Place  
the IC half way between the two electrodes to match Cx  
loading. Avoid routing ground plane (or other traces) close  
to either sense line or the electrodes; allow 4-5 mm  
clearance from any ground or other signal line to the  
electrodes or their wiring. Where ground plane is required  
(for example, under and around the QT320 itself) the  
sense wires should have minimized adjacency to ground.  
In some cases the circuit may be too sensitive, even with high  
signal threshold values. In this case gain can be lowered by  
making the electrode smaller, using sparse mesh with a high  
space-to-conductor ratio (Figure 1-3), and most importantly by  
decreasing Cs. Adding Cx capacitance will also decrease  
sensitivity.  
It is also possible to reduce sensitivity by making a capacitive  
divider with Cx by adding a low-value capacitor in series with  
the electrode wire.  
2. Connect a small capacitor (~5pF) between S1a or S1b  
(either Channel 1 pin) and circuit ground (Csx in Figure  
1-6), this will increase the load capacitance of Channel 1,  
thus balancing the sensitivity of the two channels (see  
Figures 1-7, 1-8).  
1.4.3 HYSTERESIS  
Hysteresis is required to prevent chattering of the output lines  
with weak, noisy, or slow-moving signals.  
3. Adjust Cs and/or the internal threshold of the two channels  
until the sensitivities of the two channels are  
indistinguishable from each other.  
The hysteresis can be set independently per channel.  
Hysteresis is a reference-based number; thus, a threshold of  
10 with a hysteresis of 2 will yield 2 counts of hysteresis  
(20%); the channel will become active when the signal equals  
or exceeds a count of 10, and go inactive when the count falls  
to 7 or lower.  
Since the actual burst length is proportional to sensitivity, you  
can use an oscilloscope to balance the two channels with  
more accuracy than by empirical methods (See Figures 1-7  
and 1-8). Connect one scope probe to Channel 1 and the  
other to Channel 2, via large resistors (750K ohms) to avoid  
disturbing the measurement too much, or, use a low-C FET  
probe. The Csx balance capacitor should be adjusted so that  
the burst lengths of Channels 1 and 2 look nearly the same.  
Hysteresis can also be set to zero (0), in which case the  
sensor will go inactive when the count falls to 9 or lower in the  
above example.  
Threshold levels of under 4 counts are hard to deal with as  
the hysteresis level is difficult to set properly.  
With some diligence the PCB can also be designed to include  
some ground plane nearer to Channel 1 traces to induce  
about 5pF of Csx load without requiring an actual discrete  
capacitor.  
1.4.4 CHANNEL  
B
ALANCE  
Channel 1 has less internal Cx than Channel 2, which makes  
it more sensitive than Channel 2 given equal Cx loads and Cs  
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QT320/R1.03 08/02  
Figure 1-9 Bursts when SC > 0  
1.5 TIMING  
Figure 1-11 Burst detail  
The QT320 runs two sensing bursts, one per channel, each  
acquisition cycle (Figure 1-9). The bursts are successive in  
time, with Channel 2 firing first.  
1.5.1 BURST  
S
PACING: T , SC, TBS  
I
The basic QT320 timing parameters are:  
Between acquisition bursts, the device can go into a low  
power sleep mode. The percentage of time spent in sleep  
depends on the burst spacing and the combined burst lengths  
of both channels; if the burst lengths occupy all of the sleep  
interval, no time will be spent in sleep mode and the part will  
operate at maximum power drain.  
Ti  
Tbs  
Tbd1  
Tbd2  
Tbd  
Basic timing interval  
Burst spacing  
Burst duration, Channel 1  
Burst duration, Channel 2  
Burst duration, Ch1 + Ch2  
Max On-Duration  
(1.5.1)  
(1.5.1)  
(1.5.2)  
(1.5.2)  
(1.5.2)  
(1.5.3)  
(1.5.4)  
Tmod  
Tdet  
The burst spacing is a multiple of the basic timing interval Ti;  
Ti in turn depends heavily on Vdd (see Section 2.1 and Figure  
5.7). The parameter ‘Sleep Cycles’ or SC is the user-defined  
Setup value which controls how many Ti intervals there are  
from the start of a burst on Channel 2 until the start of the  
next such burst. The resulting timing is Tbs:  
Detection response time  
Tbs = SC x Ti  
where SC > 0.  
All the basic timing parameters of the QT320 such as  
recalibration delay etc. are dependent on Tbs.  
If SC = 0, the device never sleeps between bursts (Figure  
1-10). This mode is fast but consumes maximum power; it is  
also unregulated in timing from burst to burst, depending on  
the combined burst lengths of both channels.  
Conversely if SC >> 0, the device will spend most of its time  
in sleep mode and will consume very little power, but it will be  
slower to respond.  
By selecting a supply voltage and a value for SC, it is possible  
to fine-tune the circuit for the desired speed / power tradeoff.  
1.5.2 BURST  
DURATIONS: TBD1, TBD2, TBD  
The two burst durations depend entirely on the values of Cs  
and Cx for the coresponding sensing channel, and to a lesser  
extend, Vdd. The bursts are composed of hundreds of  
charge-transfer cycles (Figure 1-11) operating at about  
500kHz. Channel 2 always fires first (Tbd2) followed by  
Channel 1 (Tbd1); the sum total of the time required by both  
channels is parameter Tbd.  
Figure 1-10 Bursts when SC = 0  
(750K resistor in series with scope probe)  
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QT320/R1.03 08/02  
When SC=0 (no sleep cycles), the sensor operates without a  
fixed timing and the acquisition spacing Tbs is the sum of the  
burst durations for both channels (Figure 1-10). In this mode  
of operation, Tbs and Tbd are the same value.  
2 - CONTROL & PROCESSING  
All acquisition functions are digitally controlled and can be  
altered via the cloning process.  
Signals are processed using 16 bit integers, using  
Quantum-pioneered algorithms specifically designed to  
provide for high survivability.  
1.5.3 MAX  
O -DURATION, TMOD  
N
The Max On-Duration is the amount of time required for a  
continuously detecting sense channel to recalibrate itself. This  
parameter is user settable by changing MOD and SC (Section  
2.6).  
2.1 SLEEP CYCLES (SC)  
Range: 0..255; Default: 1  
Affects speed & power of entire device.  
Tmod restarts if the OUT pin becomes inactive.  
A recalibration of one channel has no effect on the other;  
Tmod operates independently for each channel.  
Refer to Section 1.5.1 for more information on the effect of  
Sleep Cycles.  
1.5.4 RESPONSE  
Response time from the onset of detection to an actual OUT  
pin becoming active depends on:  
TIME, TDET  
SC changes the number of intervals Ti separating two  
consecutive burst pairs (Figure 1-10). SC = 0 disables sleep  
intervals and bursts are crowded together with a rep rate that  
depends entirely on the burst lengths of both channels  
(Section 1.5.2).  
Ti  
SC  
Basic Timing Interval  
Sleep Cycles  
DIT Detection Integrator Target (user setting)  
(user setting)  
Response time, drift compensation rate, max on-duration, and  
power consumption are all affected by this parameter. A high  
value of SC will make the sensor very low power and very  
slow.  
DIS Detect Integration Speed  
Tbd Burst duration  
(user setting)  
(if DIS is set too fast)  
Ti depends in turn on Vdd.  
If the control bit DIS is normal (0), then Tdet depends on the  
rate at which the bursts are acquiring, and the value of DIT. A  
DIT number of bursts must confirm the detection before the  
OUT line becomes active:  
2.2 DRIFT COMPENSATION (PDC, NDC)  
Signal drift can occur because of changes in Cx, Cs, Vdd,  
electrode contamination and aging effects. It is important to  
compensate for drift, otherwise false detections and sensitivity  
shifts can occur.  
Tdet = SC x Ti x DIT  
(normal DIS)  
If DIS is set to fast, then Tdet also depends on BL:  
Drift compensation is performed by making the signal’s  
reference level slowly track the raw signal while no detection  
is in effect. The rate of adjustment must be performed slowly,  
otherwise legitimate detections could be affected. The device  
compensates using a slew-rate limited change to the signal  
reference level; the threshold and hysteresis points are slaved  
to this reference.  
Tdet = (SC x Ti) + (DIT-1)*Tbd  
(fast DIS)  
Ti depends in turn on Vdd; Tbd depends on Cs and Cx for  
both channels.  
Quantum’s QT3View software calculates an estimate of  
response time based on these parameters.  
Once an object is detected, drift compensation stops since a  
legitimate signal should not cause the reference to change.  
1.6 EXTERNAL RECALIBRATION  
The QT320 has no recalibration pin; a forced recalibration is  
accomplished only when the device is powered up. However,  
supply drain is low enough that the IC can be powered from a  
logic gate or I/O pin of an MCU; driving the Vdd pin low and  
high again can serve as a forced recalibration. The source  
resistance of many CMOS gates and MCU’s are  
low enough to provide direct power without  
Positive and negative drift compensation rates (PDC, NDC)  
can be set to different values (Figure 2-1). This is invaluable  
for permitting a more rapid reference recovery after a channel  
has recalibrated while an object was present and then  
removed.  
problems. A 0.01uF minimum bypass capacitor is  
required directly across Vdd to Vss.  
Figure 2-1 Drift Compensation  
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QT320/R1.03 08/02  
If SC > 0, then PDC+1 sets the number of burst spacings,  
Tbs, that determines the interval of drift compensation, where:  
2.3 THRESHOLDS (THR1, THR2)  
Range: 1..16; Default: 6  
Affects sensitivity.  
Tbs = SC x Ti  
Example: PDC = 9,  
Tbs = 100ms  
(Section 1.5.1)  
(user setting)  
The detection threshold is set independently for each channel  
via the cloning process. Threshold is measured in terms of  
counts of signal deviation with respect to the reference level.  
Higher threshold counts equate to less sensitivity since the  
signal must travel further in order to cross the detection point.  
then  
Tpdc = (9+1) x 100ms = 1 sec.  
If SC = 0, the result is multplied by 16, and Tbd becomes the  
time basis for the compensation rate, where:  
If the signal equals or exceeds the threshold value, a  
detection can occur. The detection will end only when the  
signal become less than the hysteresis level.  
Tbd = Tbd1 + Tbd2  
Example: PDC = 5,  
Tbd = 31ms  
(Section 1.5.2)  
(user setting)  
then  
2.4 HYSTERESIS (HYS1, HYS2)  
Range: 0...16; Default: 2  
Affects detection stability.  
Tpdc = (5+1) x 31ms x 16 = 2.98 sec  
NDC operates in exactly the same way as PDC.  
The hysteresis levels are set independently for each channel  
via the cloning process. Hysteresis is measured in terms of  
counts of signal deviation below the threshold level. Higher  
values equate to more hysteresis. The channel will become  
inactive after a detection when the signal level falls below  
THRn-HYSn. Hysteresis prevents chattering of the OUT pin  
when there is noise present.  
2.2.1 POSITIVE  
D
RIFT  
COMPENSATION (PDC)  
Range: 0..255; Default: 100; 255 disables  
Ability to compensate for drift with increasing signals.  
PDC corrects the reference when the signal is drifting up.  
Every interval of time the device checks each channel for the  
need to move its reference level in the positive direction in  
accordance with signal drift. The resulting timing interval for  
this adjustment is Tpdc.  
If HYS1 or HYS2 are set to a value equal or greater than  
THR1 or THR2 respectively, the channel may malfunction.  
Hysteresis should be set to between 10% and 40% of the  
threshold value for best results.  
This value should not be set too fast, since an approaching  
finger could be compensated for partially or entirely before  
even touching the sense electrode. Tpdc is common to both  
sensing channels and cannot be independently adjusted.  
If THR1 = 10 and HYS1 = 2, the hysteresis zone will represent  
20% of the threshold level. In this example the ‘hysteresis  
zone’ is the region from 8 to 10 counts of signal level. Only  
when the signal falls back to 7 will the OUT pin become  
inactive.  
2.2.2 NEGATIVE COMPENSATION (NDC)  
Range: 0...255 Default: 2; 255 disables  
D
RIFT  
Aability to compensate for drift with decreasing signals.  
This corrects the reference level when the signal is  
decreasing due to signal drift. This should normally be faster  
than positive drift compensation in order to compensate  
quickly for the removal of a touch or obstruction from the  
electrode after a MOD  
recalibration (Section 1.5.3).  
This parameter is common to  
both channels. The resulting  
timing interval for this  
adjustment is Tndc.  
Figure 2-2 Detect Integrator Filter Operation  
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QT320/R1.03 08/02  
The MOD function can also be disabled, in which case the  
channel will never recalibrate unless the part is powered down  
and back up again. In infinite timeout the designer should  
take care to ensure that drift in Cs, Cx, and Vdd do not cause  
the device to ‘stick on’ inadvertently when the target object is  
removed from the sense field.  
2.5 DETECT INTEGRATORS (DIA, DIB, DIS)  
DIAT1, 2 Range: 1..256 Default: 10  
DIBT1, 2 Range: 1..6  
DIS Range: 0, 1  
Affects response time Tdet.  
Default: 6  
Default: 1  
See Figure 2-2 for operation.  
MOD is expressed in multiples of the burst space interval,  
which can be either Tbs or Tbd depending on the Sleep  
Cycles setting (SC).  
It is usually desirable to suppress detections generated by  
sporadic electrical noise or from quick contact with an object.  
To accomplish this, the QT320 incorporates two detection  
integrator (‘DI’) counters per channel that serve to confirm  
detections and slow down response time. The counter pairs  
operate independently for each sensing channel.  
If SC > 0, the delay is:  
Tmod = (MOD + 1) x 16 x Tbs  
Example:  
DIA / DIAT: The first counter, DIA, increments after each  
burst if the signal threshold has been exceeded in that burst,  
until DIA reaches its terminal count DIAT, after which the  
corresponding OUT pin goes active. If the signal falls below  
the threshold level prior to reaching DIAT, DIA resets.  
Tbs = 100ms,  
MOD = 9;  
Tmod = (9 + 1) x 16 x 100ms = 160 secs.  
If SC = 0, Tmod is a function of the total combined burst  
durations, Tbd. If SC = 0, the delay is:  
DIA can also be viewed as a 'consensus' filter that requires  
signal threshold crossings over ‘T’ successive bursts to create  
an output, where ‘T’ is the terminal count (DIAT).  
Tmod = (MOD + 1) x 256 x Tbd  
Example:  
DIA1 / DIAT1 and DIA2 / DIAT2 are used in conjunction with  
their respective channels.  
Tbd = 18ms,  
MOD = 9;  
DIB / DIBT: If OUT is active and the signal falls below the  
hysteresis level, detect integrator DIB, counts up towards  
terminal count DIBT; when DIBT is reached, OUT is  
deactivated. DIBT is the same as DIAT if DIBT <= 6;  
If DIAT > 6, then DIBT = 6.  
Tmod = (9 + 1) x 256 x 18ms = 46 secs.  
If MOD = 255, recalibration timeout = infinite (disabled)  
regardless of SC.  
An MOD induced recalibration will make an OUT pin inactive  
except if the output is set to toggle mode (Section 2.7.2), in  
which case the OUT state will be unaffected but the  
underlying channel will have recalibrated.  
DIBT cannot be adjusted separately from DIAT.  
DIS: Because the DI counters count at the burst rate, slow  
burst spacings can result in very long detection delays with  
terminal counts above 1. To cure this problem, the burst rate  
can be made faster while DIA or DIB is counting up. This  
creates the effect of a gear-shifted detection process: normal  
speed when there are no threshold crossings, and fast mode  
when a detection is pending. The control bit for the fast DI  
mode is referred to as DIS. DIS applies to both channels; it  
cannot be enabled for just one channel.  
2.7 OUTPUT FEATURES  
Available output processing options accommodate most  
requirements; these can be set via the clone process.  
Both OUT pins are open-drain, and require pullup resistors.  
2.7.1 DC MODE, POLARITY  
DIS gear-shifts the effect of both DIA and DIB. The  
gear-shifting ceases and normal speed resumes once the  
detection is confirmed (DIA = DIAT) and once the detection  
ceases (DIB = DIBT).  
In DC mode the OUT pins respond to detections with a  
steady-state active logic level, this state will endure for the  
length of time that a detection exists or until a MOD timeout  
occurs (Section 2.6).  
When SC=0 the device operates without any sleep cycles,  
and so the timebase for the DI counters is very fast.  
The polarity of OUT can be set via the cloning process. Each  
channel can be set for this feature independently. Either  
active-low or active-high can be selected.  
2.6 MAX ON-DURATION (MOD)  
Range: 0..255; Default: 14; 255 disables  
Affects parameter Tmod, the calibration delay time  
2.7.2 TOGGLE  
M
ODE  
Toggle mode gives OUT pins a touch-on / touch-off flip-flop  
action, so that its state changes with each detection. It is most  
useful for controlling power loads, for example kitchen  
appliances, power tools, light switches, etc.  
If a stray object remains on or near the sense electrode, the  
signal may rise enough to activate an OUT pin thus  
preventing normal operation. To provide a way around this, a  
Max On-Duration (‘MOD’) timer is provided to cause a  
channel recalibration if the activation lasts longer than the  
designated timeout, Tmod.  
MOD time-outs (Section 2.6) will recalibrate the underlying  
channel but leave the OUT state unchanged.  
OUT polarity (Section 2.7.1) has no effect when toggle mode  
is engaged. The initial state at power-up of the OUT pins in  
toggle mode is always open drain (logic high).  
The timeout applies individually per channel. If one channel is  
active for the Max On-duration interval it will recalibrate, but  
the other channel will remain unaffected.  
Each channel can be set individually for this feature.  
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QT320/R1.03 08/02  
Vdd  
8
3.2 POWER SUPPLY  
VDD  
3
5
RE1  
3.2.1 STABILITY  
S1A  
S1B  
SENSOR 1  
The QT320 derives its internal references from the power  
supply. Sensitivity shifts and timing changes will occur with  
changes in Vdd, as often happens when additional power  
supply loads are switched on or off via one of the Out pins.  
1
7
RE3  
RE4  
CS1  
OUT1  
OUT2  
OUT1  
OUT2  
6
2
RE2  
These supply shifts can induce detection ‘cycling’, whereby  
an object is detected, the load is turned on, the supply sags,  
the detection is no longer sensed, the load is turned off, the  
supply rises and the object is reacquired, ad infinitum.  
SENSOR 2  
S2A  
S2B  
CS2  
VSS  
4
Detection ‘stiction’, the opposite effect, can occur if a load is  
shed when an output is active and the signal swings are  
small: the Out pin can remain stuck even if the detected  
object is no longer near the electrode.  
Figure 3-1 ESD/EMC protection resistors  
2.7.3 HEART  
B
EAT™ OUTPUT  
3.2.2 SUPPLY  
R
EQUIREMENTS  
Both OUT pins have HeartBeat™ ‘health’ indicator pulses  
superimposed on them. Heartbeat floats both 'OUT' pins for  
approximately 15µs once before Channel 2’s burst.  
Vdd can range from 1.8 to 5.25 volts during operation, and  
2.2 to 5.25 during eeprom Setups configuration. Current drain  
will vary depending on Vdd, the chosen sleep cycles, and the  
burst lengths. Increasing Cx values will decrease power drain  
since increasing Cx loads decrease burst length (Figures 5-1,  
5-4).  
These pulses can be used to determine that the sensor is  
operating properly. The pulses are evident on an OUT line  
that is low, and appear as positive pulses.  
They are not evident on an OUT pin that is high.  
If the power supply is shared with another electronic system,  
care should be taken to assure that the supply is free of  
spikes, sags, and surges. The QT320 will track slow changes  
in Vdd if drift compensation is enabled, but it can be adversely  
affected by rapid voltage steps and spikes at the millivolt  
level.  
Heartbeat indication can be used to determine if the chip is  
operating properly. The frequency of the pulses can be used  
to determine if the IC is operating within desired limits.  
It is not possible to disable these pulses.  
Heartbeat pulses can be easily filtered by placing a suitable  
capacitor from an OUT pin to Vss, to prevent the OUT line  
from rising substantially within the 15µs pulse. For example,  
with a 10K pullup resistor, the capacitor can be 0.015µF of  
virtually any type.  
If desired, the supply can be regulated using a conventional  
low current regulator, for example CMOS LDO regulators with  
low quiescent currents, or standard 78Lxx-series 3-terminal  
regulators.  
For proper operation a 100nF (0.1uF) ceramic bypass  
capacitor should be used between Vdd and Vss; the bypass  
cap should be placed very close to the Vdd and Vss pins.  
2.7.4 OUTPUT  
D
RIVE  
C
APABILITY  
The outputs can sink up to 2mA of non-inductive current. If an  
inductive load is used, such as a small relay, the load should  
be diode-clamped to prevent damage. The current must be  
limited to 2mA max to prevent detection side effects from  
occurring, which happens when the load current creates  
voltage drops on the die and bonding wires; these small shifts  
can materially influence the signal level to cause detection  
instability.  
3.3 PCB LAYOUT  
3.3.1 GROUND  
P
LANES  
The use of ground planes around the device is encouraged  
for noise reasons, but ground should not be coupled too close  
to the four sense pins in order to reduce Cx load. Likewise,  
the traces leading from the sense pins to the electrode should  
not be placed directly over a ground plane; rather, the ground  
plane should be relieved by at least 3 times the width of the  
sense traces directly under it, with periodic thin bridges over  
the gap to provide ground continuity.  
3 CIRCUIT GUIDELINES  
3.1 SAMPLE CAPACITORS  
Cs capacitors can be virtually any plastic film or low to  
medium-K ceramic capacitor. The normal usable Cs range is  
from 1nF ~ 200nF depending on the sensitivity required;  
larger values of Cs require higher stability to ensure reliable  
sensing. Acceptable capacitor types include NPO or C0G  
ceramic, PPS film, Y5E and X7R ceramic in that order.  
3.3.2 CLONE  
P
ORT  
C
ONNECTOR  
If a cloning connector is used, place this close to the QT320  
(Figure 4-1). Placing the cloning connector far from the  
QT320 will increase the load capacitance Cx of the sensor  
and decrease sensitivity, as some of the cloning lines are  
sense lines. Long distances on these lines can also make the  
clone process more susceptible to communication errors from  
ringing and interference.  
If the design requires sensitivity matching between channels,  
it is strongly advised to use tight tolerance capacitors and to  
trim the relative sensitivities as described in Section 1.4.4.  
Cloning can be designed for production by using pads (SMT  
or through-hole) on the solder side which are connected to a  
fixture via spring loaded ATE-style ‘pogo-pins’. This eliminates  
the need for an actual connector to save cost.  
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QT320/R1.03 08/02  
3.4 ESD ISSUES  
In cases where the electrode is placed behind a dielectric  
panel, the device will usually be well protected from static  
discharge. However, even with a plastic or glass panel,  
transients can still flow into the electrode via induction, or in  
extreme cases, via dielectric breakdown. Porous materials  
may allow a spark to tunnel right through the material;  
partially conducting materials like 'pink poly' static dissipative  
plastics will conduct the ESD right to the electrode. Panel  
seams can permit discharges through edges or cracks.  
Vdd  
8
VDD  
3
5
S1A  
S1B  
SENSOR 1  
CS1  
1
7
OUT1  
OUT2  
OUT1  
OUT2  
Testing is required to reveal any problems. The QT320 has  
internal diode protection which can absorb and protect the  
device from most induced discharges, up to 20mA; the  
usefulness of the internal clamping will depend on the  
dielectric properties, panel thickness, and rise time of the  
ESD transients.  
6
2
S2A  
S2B  
SENSOR 2  
CS2  
VSS  
4
ESD protection can be enhanced with an added resistor as  
shown in Figure 3-1. Because the charge and transfer times  
of the QT320 are 1us in duration, the circuit can tolerate  
values of Re which result in an RC timeconstant of about  
200ns. The ‘C’ of the RC is the Cx load on the distant side  
from the QT320. Thus, for a Cx load of 20pF, the maximum  
Re should be 10K ohms. Larger amounts of Re will result in  
an increasingly noticeable loss of sensitivity.  
Figure 4-1 Clone interface wiring  
board has been designed with a connector to facilitate direct  
connection with the QTM300CA. The QTM300CA in turn  
connects to any PC with a serial port which can run QT3View  
software (included with the QTM300CA and available on  
Quantum’s web site).  
The connections required for cloning are shown in Figure 4-1.  
Further information on the cloning process can be found in  
the QTM300CA instruction guide. Section 3.3.2 discusses  
wiring issues associated with cloning.  
3.5 EMC ISSUES  
Electromagnetic and electrostatic susceptibility are often a  
problem with capacitive sensors. QT320 behavior under these  
conditions can be improved by adding the series-R shown in  
Figure 3-1, exactly as shown for ESD protection. The resistor  
should be placed next to the chip.  
The parameters which can be altered are shown in Table 4-1  
(next page).  
This works because the inbound RC network formed by Re  
and Cs has a very low cutoff frequency which can be  
computed by the formula:  
Parameters that can be altered for each channel  
independently are:  
Threshold  
Hysteresis  
Detect Integrator A  
Detect Integrator B  
Max On-Duration  
Output Mode  
1
Fc =  
2✜ Re Cs  
If Re = 10K and Cs = 10nF, then Fc = 1.6kHz.  
This leads to very strong suppression of external fields.  
Nevertheless, it is always wise to reduce lead lengths by  
placing the QT320 as close to the electrodes as possible.  
Parameters that are common to the entire part are:  
Detect Integrator Speed  
Negative Drift Compensation  
Positive Drift Compensation  
Sleep Cycles  
Likewise, RF emissions are sharply curtailed by the use of  
Re, which bandwidth limits RF emissions based on the value  
of Re and Cx, the electrode capacitance.  
Line conducted EMI can be reduced by making sure the  
power supply is properly bypassed to chassis ground. The  
OUT lines can also be paths for conducted EMI, and these  
can be bypassed to circuit ground with an RC filter network.  
It is possible for an on-board host controller to read and  
change the internal settings via the interface, but doing so will  
inevitably disturb the sensing process even when data  
transfers are not occuring. The additional capacitive loading  
of the interface pins will contribute to Cx; also, noise on the  
interface lines can cause erratic operation.  
4 PARAMETER CLONING  
The cloning process allows user-defined settings to be loaded  
into internal eeprom, or read back out, for development and  
production purposes.  
The internal eeprom has a life expectancy of 100,000  
erase/write cycles.  
A serial interface specification for the device can be obtained  
by contacting Quantum.  
The QTM300CA cloning board in conjunction with QT3View  
software simplifies the cloning process greatly. The E3B eval  
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TABLE 4-1 SETUPS SUMMARY CHART  
Description  
Symbol  
THR1  
Valid Values  
Default  
Calculation / Notes  
Higher = less sensitive  
Unit  
Counts  
Threshold  
1 - 16  
-
-
-
6
2
Hysteresis  
HYS1  
0 - 16  
Higher = more hysteresis  
Counts  
Det Integrator A  
DIAT1  
1 - 256  
10  
Higher = longer to detect, more noise immune  
Burst Cycles  
Det Integrator B  
Max-On Duration  
DIBT1  
MOD1  
1 - 6  
-
6
Value taken from DIAT1 but truncated to 6  
Channel 1  
Specific  
0 - 254  
255  
0
Finite  
SC = 0  
SC > 0  
Tmod = (MOD1 + 1) x 256 x Tbs (note1)  
Tmod = (MOD1 + 1) x 16 x Tbs (note2)  
14 (~10s at 3V)  
Seconds  
Infinite  
Active Low  
-
-
-
1
Active High  
Output Mode  
OUT1  
0
Requires pullup resistor on OUT1  
2
Toggle  
Threshold  
Hysteresis  
THR2  
HYS2  
DIAT2  
1 - 16  
0 - 16  
1 - 256  
-
-
-
6
2
Higher = less sensitive  
Counts  
Counts  
Higher = more hysteresis  
Det Integrator A  
10  
Higher = longer to detect, more noise immune  
Value taken from DIAT2 but truncated to 6  
Burst Cycles  
Det Integrator B  
Max-On Duration  
DIBT2  
MOD2  
1 - 6  
-
6
Channel 2  
Specific  
0 - 254  
Finite  
Infinite  
Active Low  
Active High  
Toggle  
Slow  
SC = 0  
Tmod = (MOD2 + 1) x 256 x Tbs (note1)  
Tmod = (MOD2 +1) x 16 x Tbs (note2)  
14 (~10s at 3V)  
Seconds  
255  
SC > 0  
0
-
-
-
1
2
Output Mode  
DI Speed  
OUT2  
0
1
Requires pullup resistor on OUT2  
0
-
-
-
-
DIS  
NDC  
PDC  
SC  
1
Fast  
0 - 254  
255  
0 - 254  
255  
0
On  
SC = 0  
SC > 0  
SC = 0  
SC > 0  
Tndc = (NDC + 1) x 16 x Tbs (note1)  
Negative Drift  
Compensation  
2 (~0.13s/bit  
@ 3V)  
Seconds / bit  
change  
Features  
Common  
To Both  
Off  
Tndc = (NDC + 1) x Tbs  
(note2)  
On  
Tpdc = (PDC + 1) x 16 x Tbs (note1)  
Positive Drift  
Compensation  
100 (~4.36s/bit  
@ 3V)  
Seconds / bit  
change  
Channels  
Off  
Tpdc = (PDC + 1) x Tbs  
(note2)  
No Sleep  
Sleep  
1 (~47ms Tbs  
@ 3V)  
Sleep Cycles  
Burst rep interval = Tbs = SC x Ti  
Counts  
1 - 255  
Note 1: Tbs is the combined (summed) burst duration of Channel1 and Channel2 (Tbd).  
Note 2: Tbs is variable with the voltage, see figure 5-7. If Tbd is longer than 10ms,Tbs is Tbd plus the sleep time find on figure 5-7.  
Note 5: The sleep period time is find on figure 5-7(equivalent at 1 sleep period).  
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5 ELECTRICAL SPECIFICATIONS  
5.1 ABSOLUTE MAXIMUM SPECIFICATIONS  
Operating temp. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . as designated by suffix  
Storage temp. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . -65OC to +150OC  
VDD. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . -0.5 to +6V  
Max continuous pin current, any control or drive pin. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . ±±0mꢀ  
Short circuit duration to ground, any pin. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . infinite  
Short circuit duration to VDD, any pin. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . infinite  
Voltage forced onto any pin. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . -0.5V to (Vdd + 0.5) Volts  
5.2 RECOMMENDED OPERATING CONDITIONS  
V
V
DD. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . +1.8 to 5.5V  
DD during eeprom writes. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . +2.2 to 5.5V  
Short-term supply ripple+noise. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . ±5mV  
Long-term supply stability. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . ±100mV  
Cs value. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1nF to 200nF  
Cx value. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 0 to 100pF  
5.3 AC SPECIFICATIONS  
Vdd = 3.0, Ta = recommended operating range, Cs=100nF unless noted  
Symbol  
Description  
Min  
Typ  
150  
1
Max  
Units  
ms  
µs  
Notes  
Cs, Cx dependent  
TRC  
Recalibration time  
Charge duration  
Transfer duration  
Burst length  
TPC  
T
PT  
BL  
1
µs  
T
0.5  
25  
ms  
µs  
Cs = 10nF to 200nF; Cx = 0  
THB  
Heartbeat pulse width  
15  
5.4 SIGNAL PROCESSING  
Symbol Description  
Min  
Typ  
Max  
Units  
counts  
counts  
samples  
ms/bit  
ms/bit  
secs  
Notes  
Threshold differential w.r.t. reference  
1
0
1
16  
15  
Hysteresis w.r.t. threshold  
Consensus filter length  
256  
Positive drift compensation rate  
Negative drift compensation rate  
-
-
Post-detection recalibration timer duration  
<1  
infinite  
5.5 DC SPECIFICATIONS  
Vdd = 3.0V, Cs = 10nF, Cx = 5pF, Ta = recommended range, unless otherwise noted  
Symbol  
Description  
Supply voltage  
Min  
1.8  
2.2  
60  
Typ  
Max  
5.25  
Units  
V
Notes  
V
DD  
V
V
DDW  
Vdd during eeprom write  
Supply current  
5.25  
V
I
DD  
600  
1,500  
µA  
V/s  
V
Depends on setting of Sleep Cycles  
Required for proper start-up  
Vdd = 2.5 to 5.0V  
DDS  
Supply turn-on slope  
Input low voltage  
100  
V
IL  
0.3 Vdd  
V
IH  
Input high voltage  
0.6 Vdd  
0
V
Vdd = 2.5 to 5.0V  
V
OL  
Low output voltage  
0.4  
200  
16  
V
OUT1, OUT2, 2mA sink  
C
X
Load capacitance range  
Acquisition resolution  
Sensitivity range, Channel 1  
Sensitivity range, Channel 2  
pF  
bits  
pF  
pF  
A
R
S1  
S2  
0.4  
0.6  
Threshold = 6; ref. Figure 5-3  
Threshold = 6; ref. Figure 5-4  
lQ  
13  
QT320/R1.03 08/02  
10.00  
1.00  
0.10  
0.01  
10.00  
1.00  
0.10  
0.01  
4.7nF  
9nF  
4.5nF  
9nF  
19nF  
43nF  
74nF  
124nF  
200nF  
19nF  
43nF  
74nF  
124nF  
200nF  
0
10  
20  
30  
40  
50  
0
10  
20  
30  
40  
50  
Cx Load  
Cx Load  
Figure 5-2 Typical Ch 2 Sensitivity vs. Cx;  
Threshold = 16, Vdd = 3.0  
Figure 5-1 Typical Ch 1 Sensitivity vs. Cx;  
Threshold = 16, Vdd = 3.0  
10.00  
10.00  
1.00  
0.10  
0.01  
4.7nF  
9nF  
1.00  
0.10  
0.01  
4.7nF  
9nF  
19nF  
43nF  
74nF  
124nF  
200nF  
19nF  
43nF  
74nF  
124nF  
200nF  
0
10  
20  
30  
40  
50  
0
10  
20  
30  
40  
50  
Cx Load  
Cx Load  
Figure 5-3 Typical Ch 1 Sensitivity vs. Cx;  
Threshold = 6, Vdd = 3.0  
Figure 5-4 Typical Ch 2 Sensitivity vs. Cx;  
Threshold = 6, Vdd = 3.0  
lQ  
14  
QT320/R1.03 08/02  
25.000  
20.000  
25.000  
20.000  
15.000  
10.000  
5.000  
15.000  
10.000  
5.000  
Cx = 0pF  
Cx = 0pF  
0.000  
Cx = 21pF  
0.000  
52  
Cx = 21pF  
118  
228  
Cx = 48pF  
Load (pF)  
52  
507  
884  
118  
228  
1450  
2357  
507  
Cx = 48pF  
Load (pf)  
884  
1450  
Sampling Capacitor (nF)  
2357  
Sampling Capacitor (nF)  
Figure 5-6 Typical Ch 2 burst length vs Cx, Cs;  
Vdd = 3.0  
Figure 5-5 Typical Ch 1 burst length vs Cx, Cs;  
Vdd = 3.0  
180  
160  
140  
120  
100  
80  
60  
40  
20  
0
1.5  
2
2.5  
3
3.5  
4
4.5  
5
5.5  
Power Supply (Volts)  
Figure 5-7 Typical total burst spacing vs. Vdd;  
SC = 1, Tbd < 10ms  
lQ  
15  
QT320/R1.03 08/02  
450  
400  
350  
300  
250  
200  
150  
100  
50  
Sleep Cycles  
None  
One  
Two  
Three  
Five  
0
0
10  
20  
30  
40  
50  
60  
Sampling Capacitor (nF)  
Figure 5-8 Idd current vs Cs; Vdd = 2.0  
900  
800  
700  
600  
500  
400  
300  
200  
100  
0
Sleep Cycles  
None  
One  
Two  
Three  
Five  
Ten  
0
10  
20  
30  
40  
50  
60  
Sampling Capacitor (nF)  
Figure 5-9 Idd current vs Cs; Vdd = 3.3  
2000  
1800  
1600  
1400  
1200  
1000  
800  
Sleep Cycles  
None  
One  
Two  
Three  
Five  
Ten  
600  
400  
200  
0
0
10  
20  
30  
40  
50  
60  
Sampling Capacitor (nF)  
Figure 5-10 Idd current vs Cs; Vdd = 5.0  
lQ  
16  
QT320/R1.03 08/02  
M
A
F
S1  
a
A
r
S
L2  
Pin 1  
x
m
Q
L1  
L
Package type: 8-pin Dual-In-Line  
Millimeters  
Inches  
SYMBOL  
Min  
6.1  
Max  
7.11  
8.26  
10.16  
-
Notes  
Min  
0.24  
0.3  
Max  
0.28  
0.325  
0.4  
Notes  
a
A
7.62  
9.02  
7.62  
0.69  
0.356  
1.14  
0.203  
2.54  
0.38  
2.92  
-
M
m
Q
L
0.355  
0.3  
Typical  
-
Typical  
0.94  
0.559  
1.78  
0.305  
-
0.027  
0.014  
0.045  
0.008  
0.1  
0.037  
0.022  
0.07  
0.012  
-
L1  
L2  
F
BSC  
BSC  
r
-
0.015  
0.115  
-
-
S
3.81  
5.33  
10.9  
0.15  
0.21  
0.43  
S1  
x
M
M
a
H
A
φ
e
h
Pin 1  
E
F
L
Package type: 8-pin Wide SOIC  
Millimeters  
Inches  
SYMBOL  
Min  
5.21  
Max  
5.41  
Notes  
Min  
0.205  
0.3  
Max  
0.213  
0.33  
Notes  
a
A
M
F
L
7.62  
5.16  
8.38  
5.38  
0.203  
0.05  
0.212  
1.27  
BSC  
BSC  
0.305  
0.102  
1.78  
0.508  
0.33  
2.03  
0.012  
0.004  
0.07  
0.02  
0.013  
0.08  
h
H
e
0.178  
0.508  
o
0.254  
0.889  
o
0.007  
0.02  
o
0.01  
0.035  
o
E
φ
0
8
0
8
lQ  
17  
QT320/R1.03 08/02  
lQ  
©2002 QRG Ltd.  
Patented and patents pending  
Corporate Headquarters  
1 Mitchell Point  
Ensign Way, Hamble SO31 4RF  
Great Britain  
Tel: +44 (0)23 8056 5600 Fax: +44 (0)23 8045 3939  
admin@qprox.com  
www.qprox.com  
North America  
651 Holiday Drive Bldg. 5 / 300  
Pittsburgh, PA 15220 USA  
Tel: 412-391-7367 Fax: 412-291-1015  
Specifications subject to change.  
This device expressly not for use in any medical or human safety related  
application without the express written consent of an officer of the company.  
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